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Measurement Concepts 




Significant Contributions 






PRICE $1.00 

(©TEKTRONIX, INC.; lybb 




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INDEX -^5 


The concepts for the measurement of semiconductor 
characteristics that are covered in this book 
are embodied by instruments and methods widely used 
today. The purpose of the book is not to expose new 
ideas, although we certainly hope some will be new 
to the average reader. Instead, its purpose is to 
corral what we believe to be many of the better 
ideas already in use, and discuss them, particularly 
ideas that involve the use of Tektronix instruments. 

The reasons for measuring the characteristics of 
semiconductor devices fall into about four categories. 
Either the measurement is for the purpose of producing 
better components, sorting components, predicting 
performance in a circuit, or improving a circuit. 
The characteristics of semiconductor devices that 
are of practical importance to their use in an 
electrical circuit can usually be measured with an 
electrical instrument. Many of those measurements 
also provide good analytical information for people 
improving component design or maintaining production 
quality and specifications. 

Much of the discussion relates to measurements 
actually performed, using specific semiconductor types 
and instrument types. These will exemplify a variety 
of measurement considerations and concepts. Only 
measurements on discrete semiconductor components 
are discussed. Integrated circuits are not covered. 

We hope the book may help engineers and technicians 
make more meaningful and accurate tests and 
measurements of the characteristics of diodes, 
transistors, and other semiconductor devices. 



Bipolar transistors are those transistors which 
normally use current carriers of both polarities. 
The category consists mainly of the familiar three- 
terminal two-j unction, NPN or PNP types made of 
either germanium or silicon. 


^FE — Static Forward Current Transfer Ratio 
(Common Emitter) 

^FE, The static forward current transfer ratio, 7z FE , of a 
DC beta transistor, otherwise known as DC beta or DC current 
DC current gain is simply the ratio of its collector current 
9 a ' n to its base current, assuming, of course, that the 
polarity and magnitude of the applied currents and 
voltages are within what could be called a correct, 
normal operating range for the transistor. The 
current gain of any particular transistor is apt 
to vary considerably, depending on where within its 
normal operating range the transistor is operating. 
Therefore, to be more specific when we refer to the 
static forward-current transfer ratio of a transistor, 
we should say what the collector voltage is supposed 
to be, and what either the base current or collector 
collector current is supposed to be. Usually the collector 
current current is specified, so the base current must be 
or base varied until the specified collector current flows. 
current Sometimes the base current may be specified. In that 
specified case the resultant collector current is then measured. 





B f\ 






Fig. 1-1. ?z FE , 2N39U4 50-uA base-current step. 

The measurement of base current, collector current, 
and collector voltage may be done with DC meters to 
determine DC beta. This method soon runs into a 
power problem when the power dissipated is enough to dominate 
dissipation the temperature of the transistor. It may even cause 
may limit the transistor to be burned out before the test is 
measurements completed. It is fundamental that any test or 

measurement performed on a semiconductor should not 
alter its characteristics. Care should always be 
taken to avoid applying excessive currents or 
voltages, or both, or the characteristics may be 
altered. Interrupting the applied currents and 
voltages frequently may be required to keep the 
internal temperature down, close to that of the 
desired surrounding temperature. 

A continually-pulsed mode of measuring DC beta 
suggests itself for conditions that might otherwise 
limit the accuracy of the results due to a change 
continually- in temperature. For determining DC beta under a 
pulsed mode wide variety of operating conditions few methods 
for DC beta can do better than plotting one or more curves 

to show the entire forward transfer characteristics 
for the particular transistor being used in the 
measurement. Transistor curve tracers do that 

Fig. 1-1 shows the collector current which flows as 
the collector voltage of a transistor is swept between 
zero volts and ten volts by a fullwave rectified 
sinewave after a specific base current of 50 yA has 
been applied. The DC beta can be determined for any 
point on the curve by reading the collector current 
at that point from the calibrated vertical scale, 
then dividing that current by the base current 
selected. An alternate way is to first determine what 
the beta per division is for the vertical scale, 
DC beta and read DC beta directly from the scale. For 
a + any example, the beta per division in Fig. 1-1 is 20, 
P° in+ the quotient of luA (collector current per division) 
and 50 uA (the base current per step) . 

If the purpose in measuring the DC beta of the 
transistor shown in Fig. 1-1 was to see whether it 
exceeded 180 when the collector voltage was 
5 volts and the base current was 50 pA, the 
measurement would consist simply of observing whether 
the curve was above or below Point A, the ninth 
division at the center vertical graticule line, 


qo, no-go 


Fig. 1-2. fr FE , 2N3904 53-uA base-current step 
(50 uA plus 3 uA of offset) . 

because the ninth division corresponds to a beta of 
180, and that line corresponds to 5 volts. Such a 
measurement may also be considered a test; a 
quantitative test. Rapid go, no-go testing may be 
performed without ever making a reading from the 
scale or recording a number. 

Fig. 1-2 shows the same transistor passing a 
collector current of precisely 9 mA at 5 volts, 
slightly more than in Fig. 1-1. This collector 
current was achieved by increasing the base current 
slightly. To determine the DC beta under this set 
of conditions the collector current should be 
divided by the base current as before. The only 
difference is that DC beta would now be determined 
at a specific collector current rather than at a 
specific base current. 



i nf I uence; 




Measuring DC beta at high currents or high collector 
voltages in the foregoing way may increase the 
temperature enough to influence the validity of the 
measurement. The temperature may be considerably 
reduced by reducing the percentage of time the 
transistor is turned on. This may be achieved in a 
couple of ways. One way is to plot a single-shot 
curve (single family) . A push button or lever 
switch may be provided for this purpose, and applies 
base current for only one half of one complete 
alternation of the power line per curve, 8.3 ms or 
10 ms depending on whether the power line frequency 
is 60 Hz or 50 Hz. Base current drive is removed 
except momentarily when the pushbutton is depressed. 
See Fig. 1-3. 

V c (VOLTS) 










Fig. 1-3. /?pe, 2N3904 single-shot curve. 
/zpE = 5.8, when Vq = 2.5 volts. 

C.11 UU.UW ^v*i. fc w..- — — U 

very short 
duty factor 

••n — j o o 

collector current to flow for l/120th of a second 
is sometimes long enough to burn out a small 
transistor, particularly when collector voltage is 
also high. To determine current gain for 
exceptionally high collector voltages or collector 
currents requires turning the transistor on for 
even shorter intervals of time. Fortunately this 
can usually be done repetitively at a very low duty 
factor, so that the average dissipation is low. 
Pulses of specific amounts of base current may be 
introduced for intervals as low as 80 us at a 
repetition rate as low as 50 or 60 Hz on the 
Type 576 Tektronix Curve Tracer. 

See Fig. 1-4. Peak power delivered under these 
conditions is more than 200 times greater than the 
average power. 



Fig. 1-4. h-p£, 2N3904 repetitively pulsed while 
manually scanned once. 

the base 

When this method is used the base current may be 
pulsed at moments when the swept collector voltage 
is near its peak value. Or a DC voltage, which may 
be manually varied, may be applied to the collector. 
Either way the transistor conducts only a small 
percentage of the time, at moments when base current 
pulses are applied. The collector voltage may be 
set to plot DC beta at a particular collector 
voltage, or it may be manually varied to simulate 
a curve. Fig. 1-4 is a time exposure showing the 
whole range of collector voltages below 5 volts as the 
collector voltage supply is varied with the peak- 
collector-volts knob. 






rises unti I 
heat out = 

heat in 

Sometimes the main problem with dissipating heat when 
measuring the characteristics of a transistor is 
knowing that you have a problem! Usually with a 
transistor curve tracer you may determine when heat 
dissipation becomes significant by simulating the 
measurements at lower currents and voltages, and 
increasing drive until the effects of heat become 
apparent. This procedure may require operating the 
transistor at higher collector voltages or with 
greater collector current than the test or 
measurement calls for before the effects are noticed. 
When it does, of course, the conditions for the 
desired measurement do not involve a significant 
heat effect. Reasonable care should be taken to 
not exceed collector breakdown voltage, or to use a 
resistor in series with the collector supply that 
has a high enough value to limit collector current 
to a safe value if breakdown is exceeded. Should 
the effects of excessive heat become apparent when 
even less power is dissipated than required for the 
measurement, the method of making the measurement 
will usually need to be changed. 

Whenever peak collector voltage is high, more heat 
is produced. How rapidly heat may be dissipated from 
a transistor will depend on the transistor construction 
and what method, if any, is used to transfer the heat 
away. The first hurdle in getting heat out of a 
transistor is transferring the heat developed in the 
semiconductor material to the case of the transistor. 
Temperature will build up rapidly in a transistor 
whenever a larger amount of heat is generated than 
can escape rapidly. Temperature invariably increases 
until the heat escapes at the same rate it is being 

Knowing how to recognize symptoms of excessive heat 
is important. Probably the best procedure to toiiow 
is to increase the collector sweep voltage slowly 
overheat while observing the resulting curves. If at any 
symptoms time while increasing the maximum collector voltage, 
any curve is not simply an extension of the curve 
depicted with less peak, collector voltage applied, 
there is probably excessive heat. Usually when an 
increase in temperature becomes significant, the 
curve will shift toward a different set of collector- 
current values. This can be observed quite readily 
while varying the peak collector voltage slowly. 
See Fig. 1-5. This photograph is a double exposure 
showing a repetitively swept peak collector voltage 
of 2.5 volts for the top curve and 5 volts for the 
lower curve. Notice also the prominent loop in the 
longer curve. This loop indicates a significant 
change in junction temperature during the time of each 


B /V 

— W 








Fig. 1-5. TzpE, 2N3904 effects of temperature. 

sweep. It is apparent in Fig. 1-3 also, even though 
the average temperature of the transistor is much 
less in Fig. 1-3 than in Fig. 1-5. 

Fig. 1-6 shows the same transistor being tested as 
shown in Fig. 1-5, except less heat is being generated. 
Reduced base current drive, and consequently reduced 
collector current, account for reduced heat. Also 
peak collector voltage was reduced from 2.5 volts to 
1 volt. Notice how the longer curve is close to 
being a simple extension of the shorter curve, 
compared to Fig. 1-5. 

When it is necessary to measure the DC beta of a 
transistor under conditions where temperature is 
affecting the measurement, there are only two 
approaches to the problem — either reduce the energy 
input to the transistor, or get rid of the heat 
faster. Both may sometimes be necessary. 









Fig. 1-6. h ¥E , 2N3904 effects of reduced 
temperature . 


T rise 
fast with 
low mass 

The first thing that may normally be done to reduce 
the heat generated under test conditions used for 
Fig. 1-5 and Fig. 1-6 is to reduce the repetition 
rate of the base current step generator from 50 or 
60 Hz to a one-shot basis. This way, only once each 
time a pushbutton or lever switch is activated, is 
the selected base current applied. The base current 
pulse would be applied for only half of the period 
of one cycle of the line frequency — usually 8.3 ms 
or 10 ms. Large power transistors do not normally 
change temperature appreciably in 10 ms, so one-shot 
testing of them is generally a satisfactory method. 
Low-power, low-mass transistors may change temperature 
significantly during 10 ms. When this happens the 
curve will consist of a loop instead of a single trace. 
See Fig. 1-3 and 1-7 for a comparison of two 
transistors rated for different power. Both are 

+ rui 




V01 TAfiF 








Fig. 1-7. hyc, 2N3441 reduced temperature change 
for power transistor at comparable 
peak power. 


dissipating about the same peak power. Notice the 
temperature distinct loop for Fig. 1-3. The loop is caused by 
hysteresis junction temperature rising as collector voltage 

rises towards a peak value. As the collector sweep 
voltage drops back down from its peak value, the 
transistor is hotter than while the collector voltage 
was rising. The retrace does not coincide with the 
forward trace because less collector current passes 
when the transistor is hotter — all other conditions 
being equal. The case of a transistor does not have 
it's hotter to be warm to the touch for the internal semiconductor 
than you material to momentarily reach a high temperature, 
think Poor thermal bonds may be detected this way by 

comparison with a similar transistor that has a good 
thermal bond. 

To measure DC beta of low power transistors where 
very high peak power is generated, we must reduce 
the time intervals during which the transistor is 
conducting to much less than 10 ms. Fig. 1-4 shows 
a plot of the DC beta of the same transistor as used 
in Fig. 1-3. The photograph is a time exposure 
produced by slowly reducing the collector voltage 
to zero while equal value base current pulses 300 ms 
wide are applied. 

Some curve tracers will not display as few as two 
curves at a time for a transistor, the way the curves 
have been illustrated. These illustrations show a 
principal curve depicting a range of collector currents 
at a selected base current and an incidental curve 
showing the collector current resulting from zero base 
current. The Tektronix Type 575 Curve Tracer shows a 
minimum of five curves, under similar conditions, one 
depicting collector current for zero base current, 
plus four others depicting collector current for 
discrete amounts of base current. To measure DC 
beta with displays of this kind you need only to 
ignore all curves but the one corresponding to the 
base current of interest. The base current of interest 
corresponding to any one of the curves is determined 
by multiplying the selected base current per step 
(amount of current increase per step) times the 
number of steps required to produce that curve. 
For example, if we were interested in the fifth 
curve (fourth above the zero base current curve) , 
and the base current increase per step was 20 mA, 
we would mentally multiply 4 x 20 to determine that 
80 mA of base current was applied when that curve 


+ r^ 

+ (WY\ 


■ c 






Fig. 1-8. fcpE, MPS918 DC beta from a family of 
curves. Four steps at 20mA per step. 
High temperature may reduce collector 
current . 

was produced. See Fig. 1-8, The curve corresponding 
to zero base current is usually a straight line and 
sometimes not considered a curve. The number of steps 
from zero is the correct number to multiply by the 
current per step. The reduction in beta due to 
increased temperature is apparent because of the 
negative slope of the top curve. 

When more than one curve is displayed other than the 
zero-current curve, we say a family of curves is 
displayed. If a set of curves is displayed only 
once, we say it is a single family display. 
Sometimes a single family must be displayed to look 
at a particular curve. 


fami ly 

static may 
not mean 


time and 

not static 

To limit temperature rise as much as possible with 
a single family display, the curves in the family 
should be as few as possible, and the curve of 
interest should be the one depicting the highest 
current. If the top curve cannot be selected to 
represent the desired base current, then a curve 
which can be made to represent the desired base 
current should be selected, and it should be as 
close to the top as possible. 

Some people will sense a sort of dilemma when they 
consider the need to limit the temperature rise of 
a transistor while measuring its static forward 
current transfer ratio. If the measurement is 
conducted to predict how a transistor will operate 
in a circuit under static conditions which are such 
that transistor temperature is bound to rise 
considerably, we should see that the term "static 
forward current transfer ratio" is sometimes a 
misnomer. Measurements of static characteristics 
on a repetitive transient basis may fail to predict 
that the transistor could behave differently or even 
burn out if operated at a higher duty factor or for 
longer conduction intervals than used in the 
measurement technique. For this reason it is 
sometimes important to know the conduction time and 
duty factor used in the measurement. The duty 
factor is determined by dividing the conduction 
time in each cycle by the time interval of each 
cycle. That decimal fraction is then multiplied 
by 100 to express the answer as a percentage. 

The measurement of the static forward-current transfer 
ratio, or the DC beta, of a transistor may be thought 
of in slightly different terms for greater clarity. 
Transistors are not used as static components; they 
are generally used for their ability to change 
current flow. What we really want to know when we 
measure the "static" forward current transfer ratio 
of a transistor is either 1) how much current the 
collector can deliver at any given collector voltage 
with a particular amount of base current, or 2) how 
much base current drive it takes for the collector 
to deliver a particular amount of current at a given 
collector voltage. Naturally the more peak power 
you want out of a transistor, the more limited its 
conduction duty factor has to be. The smaller the 
transistor is the more we have to tolerate shorter 
conduction intervals as well as limited duty factor. 


4 — Static Forward Current Transfer Ratio 
FE(INV) (collector And Emitter Leads Reversed) 


termi na I s 

Some transistors may be operated with the collector 
and emitter leads interchanged. When this is done 
the base-collector junction is forward biased, and 
the base-emitter junction is reverse biased. Most 
circuit designs do not deliberately use this mode 
of operating a transistor. However, a difference 
in characteristics in the saturation region is 
sometimes favorable for circuit design considerations. 
Typically DC beta is not as high in the reverse 



2 4 6 8 10 



->w\ — i 

;, r TTJ~L 



B / 

,u \ 

■ E 




Fig. 1-9. %E(INV). 2N3904. 



Transistors made to have very similar characteristics 
when the collector and emitter terminals are reversed 
are sometimes called bi-directional transistors, or 
symmetrical transistors. Fig. 1-9 shows a plot of 
^FE(INV) f°r the same transistor as used in Fig. 1-1. 
A comparison of the two figures will show that for 
all collector voltages, Ve(INV) is much less than 
the value of h-p-g- At point A the beta is 2. All of 
the measurement techniques and considerations that 
apply when measuring h^ may be used for measuring 

h r _ — Small-Signal Forward Current 


Transfer Ratio (Common Emitter) 

most The small-signal short-circuit forward current transfer 
common ratio, AC beta, or current gain of a transistor for 
parameter small input signals of low frequency, is probably 
hf e the most common transistor characteristic for which 

there is use and concern. It is the characteristic 
that lets us predict voltage gain or power gain in 
some circuits. As with DC beta, AC beta depends 
on where within the normal operating range the 
measurement is made. Therefore, measurements of 
small signal current gain should be made under 
specified conditions. Collector voltage should 
be known and either average base current or average 
collector current also known. 

Measurements of AC beta should always be made at the 
specified collector voltage, even though small 
percentage deviations in collector voltage typically 
measure produce extremely small errors. An expression of the 
AC beta need to measure output signal current at only the 
output AC specified collector voltage is made when we say that 
short; the output must be AC short-circuited. That is 
output Z another way of saying the output impedance must be 
low very low as far as the output signal is concerned. 

Otherwise, the collector voltage will be altered 
by the changes in collector current, and thereby 
reduce the changes in current. 

^f e : Implicit in the term "small signal forward current 

s j qnai transfer ratio" is the idea of signal amplification. 

amplification And most sma11 test signals are sinusoidal, so we 
are often lead to conclude that a measured amount 
of sinusoidal signal current must be applied, and 
the resulting sinusoidal output signal current must 
be extracted and measured to determine the current 









Fig. 1-10. Transistor curve tracer block diagram. 


transfer ratio, or current gain. Although there is 
nothing wrong with such a technique, there are 
other ways to make the measurement that are 
sometimes more practical. 

When we apply an alternating input signal current 
we simply add to, then subtract from, the base bias 
current that is already applied; there can be only 
one amount of base current at any instant. So it 
ratio of is a change or difference in base current that we 
changes are inducing when we apply a signal current and it 
is the resulting change or difference in collector 
current that we need to measure. Knowing the change 
in base current and the resulting change in collector 
current we can determine the transfer ratio, whether 
the changes are sinusoidal or some other shape as 
long as the collector voltage is known, and the rate 
of change slow enough so the high frequency 
limitations don't start to take effect. 

Transistor curve tracers introduce changes in base 
current in the form of equal-value steps; steps of 
selectable known amounts. These steps occur at the 
same rate as the collector supply voltage is swept 
between zero volts and some peak value and back to 
zero, producing a separate curve corresponding to 
each different value of base current. See Fig. 1-10 
for a functional block diagram of a transistor curve 
tracer. When the curves which are plotted depict 
collector current vs collector voltage for different 
values of base current, the change in collector 
current induced by one step of base current will be 
proportional to the vertical distance between 
adjacent curves, and can be read directly from the 
scale. Which vertical line is chosen for the scale 
will depend on what collector voltage was specified, 
because each vertical line corresponds to a 
particular collector voltage. (However, the whole 
display can be positioned a particular amount when 
desired to make a particular collector voltage 
appear on a line having small graduations.) 

Measurement of the small-signal short-circuit forward 
current transfer ratio of a transistor using a 
transistor curve tracer consists of the following 

1. Choose the vertical line corresponding to the 
specified collector voltage; 


2. Note on that line the distance between the 
two curves which appear above and below the 
specified base current (or specified 
collector current) and lie adjacent to the 
reading specified current; 


3. Translate that distance to the difference in 
collector current according to the current 
per division of the scale; 

4. Divide that collector current difference by 
the base current difference that caused it, 
depending on the current per step. 

An alternate way is to first divide the collector 
current per division by the base current per step to 
determine the beta per division. For example, base 
current steps of 1 mA per step, that produced curves 
1 division apart when the collector current per 
division was 100 mA., would indicate a small-signal 
current gain of 100. Under similar conditions, if 
the distance between curves was 1.4 divisions, the 
current gain would be 140. The Tektronix Type 576 
Curve Tracer will indicate the beta per division of 
the vertical scale so that it doesn't have to be 

When measuring small-signal short-circuit current 
transfer ratio in the way just described, attention 
"small should be given to the size of the small signal. A 
signal" reason for distinguishing between small-signal AC 
beta and large-signal AC beta is that current gain 
is sometimes different for small signals than it is 
for large signals. How small is a small signal? 
When we carefully define a small signal using numbers, 
the definition is somewhat arbitrary. In general, 
constant however, a small signal is one which is associated 
current with a current gain that is essentially constant for 
n a ]r, all smaller signals. That also tells us current gain 

for large-signals depends on the size of the signals. 

The same thing which causes a difference between 
current gain for large signals and current gain for 
small signals causes a change in gain for small 
signals i f we change the base bias current. In other 
words, a given difference in base current, starting 
with high base current, may not cause the same 
change in collector current as if that change in 
base current were made starting with low base current. 


When there is a considerable difference in current 
gain, the effect is readily apparent on a transistor 
curve tracer by the difference in vertical distance 
between the curves. Regions where the curves are 
closer together are regions of lower current gain. 
See Figs. 1-11 and 1-12. 

Typically, as shown in Figs. 1-11 and 1-12, 
transistors have lower current gain at high values 
of collector current (and very low values of 
collector current) than for medium values. 
Fortunately transistors don't usually have to be 
operated at high values of average collector current 
when they are only called upon to handle small 




+ (TO 









Fig. 1-11. Beta nonlinearity MPS918, 
0.1 mA per step. 


7" = 1 flmA 

i i nedi 
nonl inoar 




Fig. 1-12. Beta nonlinearity MPS918, 
1 mA per step. 

Another, but similar, reason for distinguishing 
between large-signal current gain and small-signal 
current gain is to distinguish between a linear and a 
nonlinear range of operation. A nonlinear range can 
cause signal distortion. AC methods of measuring AC 
beta may obscure distortion-causing nonlinearities 
that can be revealed by other methods of testing. The 
reason is that when sinusoidal signal currents are 
applied to the base of a transistor, the change in 
base current is both an increase and a decrease from 
the average or quiescent base current amount. A 
condition where an increase in base current produces 
less of a change in collector current than an equal 
decrease in base current will cause a distorted output 


waveform. But the amplitude of the waveform may not 
change appreciably. When that happens it is because 
the reduced current gain for the increasing half of 
the base current signal was nearly matched by the 
increase in gain during the other half cycle of the 
base current signal. DC methods of measuring small- 
signal current transfer ratio are sometimes superior. 
See Fig. 1-13. Here we can readily notice a 
difference in the vertical distance between curves at 
the vertical centerline signifying a change in beta 
for each of the base current steps. Consider a 
sinusoidal base current signal superimposed on a base 
bias current of 6 mA corresponding to curve B. If its 
peak amplitude were equal to one base-current step 
(2 mA) the peak-to-peak collector-current swing at the 





B /n 


♦ E 

— w\ 



Fig. 1-13. Beta, h fe 


centerline would be 3.0 divisions. Doubling the base 
current to a peak value equal to two steps would 
double the collector current swing to 6.0 divisions. 
Because 6.0 is precisely equal to twice 3.0, we would 
normally assume that the AC beta was constant for 
input signals having a peak amplitude of 4 mA or less. 
The curve tracer reveals a measurable change in beta 
for signals only half that size. For example, the 
distance between curve B and curve A is 1.4 divisions, 
whereas the distance between curve B and curve C is 
1.6 divisions. The beta per division is 20 so the 
small signal beta for the two steps is 28 and 32 
respectively. The small signal beta correctly 
measured by AC methods would have been half way 
between the two or 30. 

Another way to measure the forward current transfer 
b aS e ratio using a curve tracer is to plot base current 
current against collector current using base current as one 
versus of the coordinates. There can be a problem with this 
col lector method in knowing what part of the curves correspond 
current to a given collector voltage. The peak collector 
voltage varies from sweep to sweep, because of the 
inevitable IR voltage drop caused by collector 
current and the combined resistance of the 
collector-sweep supply and the collector-current- 
sensing resistors in series with the supply. 

h — Small-Signal Short-Cirouit Forward Current 
fb Transfer Ratio (Common Base) 

In most cases the measurement of alpha, the forward 
current transfer ratio of transistors operated in 
the common base mode, can be done more accurately by 
making the measurement in the common- emitter mode, 
and converting the answer by formula to the common- 
base mode. 


n . 
H fh = te 

1 + h r 

A beta of 10 equals an alpha of 0.91; 

A beta of 20 equals an alpha of 0.95; 

* i , _. . r nn n-nnlr in ril^ha nf , 98 ! 

A beta of 100 equals an alpha of 0.99. 



hard to 

J C 





Fig. 1-14. Alpha," h ¥B . 

A two-to-one change in beta from 50 to 100 corresponds 
to only a one per cent change In alpha. This says we 
have a very tough measurement to make with precision 
if we wish to measure alpha directly. Nonetheless, 
Fig. 1-14 shows a family of curves depicting 
collector current versus collector voltage with the 
current-step generator driving the emitter. The 
display illustrates how nearly equal to emitter 
current the collector current is for an average 
transistor. Each current step is equal to the current 
per division of the vertical scale. The top curve 
falls short of being coincident with the top of the 
scale by 3%, indicating a DC alpha of 0.97 if we 
assume that the instrument is perfect. A small error 
in the accuracy of the instrument could account for a 
gross error in measurement. Any deviation in the 
small-signal alpha from large-signal alpha is nearly 


impossible to discern. The extreme equality of 
separation between the curves suggests that grounded- 
base operation is capable of providing very linear 
output voltage swing. 

h — Small-Signal, Forward Current 
fc Transfer Ratio (Common Collector) 

The measurement of this parameter is conducted by 
first measuring beta under specified conditions, and 
adding one (1) to the answer. The common-collector 
mode is similar to the emitter-follower configuration 
where the emitter is the output terminal. Since 
emitter current is always the sum of the base current 
and the collector current, the change in emitter 
current which accompanies a change in base current is 
the base current plus the collector current. Because 
beta is the ratio of collector current to base 
current, a unit change in base current causes beta 
times that much change in collector current. 

fc J B J B 


y , „ __ Colleotor-to-Emitter Saturation Voltage, DC 

A transistor biased normally and operating in the 
common-emitter mode is said to be in saturation when 
there is too little collector voltage applied (or 
remaining) for an increase in base current to cause a 
significant increase in collector current. On a 
graph of a transistor showing collector current versus 
collector voltage for a particular base current, the 
saturation Voltage is the collector voltage at a 
point near or below the knee. On a graph showing a 
saturation family of such curves the saturation region is an area 
reason of low current and voltage below the knee of each 

curve. See Fig. 1-15. From this family of curves we 
can see that the knees of the curves occur at 
practically the same collector voltage for different 



7 C (VOLTS)- 




Fig. 1-15. Saturation region. 



B (Y 

• C 

— A/W- 



— A/V 


Fig. 1-16. Saturation region. 

CE (sat) 
at low 

Fig. 1-16 shows the saturation characteristics of a 
different type of transistor using identical test 
conditions and scale factors. Notice the lower 
saturation voltages for this transistor than for the 
one whose characteristics are graphed in Fig. 1-15. 

Measurement of collector-to-emitter saturation voltage 
at low power can be done quite readily using a 
transistor curve tracer with the kind of display shown 
in Figs. 1-15 and 1-16. It is well to remember, 
however, that both the base current and collector 
current should be specified to identify where the 
measurement should be made, if the purpose of the 
measurement is to verify a specification. If such a 
measurement should happen to be at a point on a curve 
above the knee, it will be of no consequence if the 


collector voltage is not excessive because saturation 
voltage is usually specified to be equal to or less 
than some maximum value. 

Probably the most important reason for knowing about 
saturation voltage is to predict the performance of a 
transistor used in DC-to-AC power inverters, chopping 
circuits, and logic circuits. In these applications 
it is important to know how low the collector voltage 
goes because at such times as the transistor is passing 
the most current for the longest periods of time, and 
power dissipation at the collector can quickly become 
excessive. A ten per cent reduction in saturation 
voltage can reduce collector dissipation by a 
comparable percentage. This could allow us to deliver 
extra power to the load — which may be many times 
the power dissipated by the transistor. 

^CE (sat) -- Collector-To-Emitter Saturation 

v Saturation resistance is an expression for the 

CE(sat) quotient of collector voltage (Vq) divided by 

= V C^ T C collector current (_Z" C ) for any given value of base 

current in the collector saturation region of a 
transistor operated in the common-emitter mode. 
This ratio is nearly constant for some transistors 
over a large range of collector current values. 
When it is fairly constant, collector saturation 
voltage can be estimated quite accurately over the 
same range at the given base current or higher 
base currents. A constant saturation resistance 
would appear as a curve with a straight slope which 
intersected the zero voltage and current points on 
the graph. The steepness of the slope would be a 
function of the x and y coordinates of the scale 
and the amount of saturation resistance. The term 
saturation resistance is sometimes used to mean the 
dynamic resistance, or slope, at a specified voltage 
or current point. 

Specifications of saturation resistance are usually 
for maximum tolerable values. Most measurements 
of saturation resistance are for the purpose of 
determining saturation voltage. With a display of 
a family of transistor curves showing collector 
current versus collector voltage for various base 
currents, saturation voltage can be more easily 


m « a =ur«»d riirectlv for nearly any combination of 
circumstances. So there is seldom need to calculate 
the saturation resistance except to verify a 

Fig 1-15 shows collector voltage to be 0.3 at an 
emitter current of 3 mA and a base current of 50 uA. 
The saturation resistance at that point is VqI lc - 
0.3/. 003 = 100ft. Higher base currents would show 
less' collector voltage needed at the same collector 
current, so the saturation resistance will be less 
at higher base currents. At lower values of 
collector current at the given base current, saturation 
resistance will be higher even though saturation 
voltage at lower collector currents is always less. 
Fig- 1-16 shows a different kind of saturation region, 
with values for saturation resistance that differ 
more widely than in Fig. 1-15. 

When testing or measuring the saturation voltage of 
a transistor at very high currents, pulse testing 
must be used to minimize heat dissipation. Methods 
may be employed similar to those discussed for 
measuring the static forward current transfer ratio. 


I rR0 and F fBR x CB0 — Collector-To-Base Current 
CBO tBKjLBU Leakage An d voltage Breakdown, 

Emitter Open 

The measurement of reverse bias leakage current and 
breakdown voltage can logically be considered at the 
same time. The current that flows through the 
collector-base junction when reverse biased, and 
when the emitter lead is open, is very much the same 
kind of phenomenon as occurs in a simple diode. 
Usually thaL current is relatively small, until the 
voltage is increased sufficiently and breakdown starts 
F (RR)CB0 to occur. Although the breakdown region is relatively 
made a I abrupt for most transistor junctions, a precise 
specified measurement of breakdown voltage can only be made 
temperature at a specific reverse current and at a specific 
a „/,ovor CO temperature. For the same reason reverse currents 
current should always be measured at specified reverse 

voltages and temperatures or the measurement is of 
limited use. 


1 — • — vw 





Fig. 1-17. T CB0 and ^ B R)CBO> 2N 918. Breakdown 
region between approximately 50 V 
and 70 V. 

A transistor curve tracer Is a simple and accurate 
instrument with which to plot and measure small reverse 
currents at any voltage up to the breakdown region. 
The breakdown region is identified and explored at 
the same time as reverse current is monitored, when 
desired. A primary concern with measuring breakdown 
voltage — or measuring leakage current near the 
breakdown region — is that of destroying the junction 
in the process. Transistor curve tracers allow 
insertion of resistors having high resistance values 
in a series with a swept voltage supply to limit reverse 
current to a safe value. The peak amplitude of the 
swept supply voltage can be controlled as needed. 
Fig. 1-17 shows the collector-base leakage and 
breakdown region of a typical low power silicon 


Fig. 1-18. I CB0 and F CBR]C BO> 2N918 - Ahru ^ 
breakdown region between 60 volts 
and 62 volts. Peak current allowed 
was 100uA. 

transistor. The breakdown region does not appear 
abrupt in this photograph, but the breakdown voltage 
is obviously between 50 and 70 volts. Another 
transistor of the same type is shown in Fig. 1-18 
that has a much more abrupt breakdown region. 

Fig. 1-17 and Fig. 1-18 represent measurements 
conducted at room temperature with no effort to control 
the temperature of the transistors. Reverse current 
was limited as much as possible to make the needed 
measurements. If we wish to see th<=> offers of 
increased temperature on the measurement, we can 
generate the needed heat very conveniently by slowly 


Fig. 1-19. 

J CB0 and 


2N918. Identical 

transistor L as used in Fig. 1-18 
temperature increased by increasing 
peak leakage current 100X to 10mA. 
Breakdown voltage increased from 
62 volts to 70 volts due to 
temperature rise. 

increasing the peak reverse current with the curve 
tracer. Fig. 1-19 is the same as Fig. 1-18 but 
with a peak current 10 raA instead of 0.1 mA. The 
transistor was too hot to hold. The breakdown 
voltage increased from 62 volts to 70 volts. The 
case temperature could have been monitored with a 
thermocouple attached if we had wished to make a 
measurement of its characteristics at a specific 
case temperature. The junction temperature will 
always be somewhat higher than the case temperature. 


^EBO and 7 (BR)EB0 

Emitter-To-tiase reverse ^ui-rvnu, 
And Voltage Breakdown, 
Collector Open 

J CE0 and 

^(BFOCEO are 







The measurement of reverse current and voltage 
breakdown between the emitter and base of a transistor 
when the collector terminal is left disconnected is 
performed in the same way as for the measurement of 
leakage and breakdown between the collector and base, 
already discussed. A comparison of the reverse- 
current and voltage breakdown characteristics of the 
collector-base junction of a transistor with the 
corresponding characteristics of the emitter-base 
junction of that transistor is sometimes interesting. 
The breakdown region is often a lower reverse voltage 
for the emitter-base junction than for the collector- 
base junction. 

J CE0 and 7 (B r)ceo ~ Collector Cutoff Current And 
ut,u v Voltage Breakdown, Base Open 

The reverse-current characteristics and voltage- 
breakdown characteristics of either the collector- 
base junction or the emitter-base junction of a 
transistor with the remaining terminal disconnected 
is really a measurement of a diode characteristic 
rather than a measurement of the transistor 
characteristic. Both of those measurements are useful 
for predicting some limitations of the transistor, but 
the measurement of similar characteristics involving 
conduction through both the collector-base and 
emitter-base junctions are probably more meaningful. 
The open-base condition, while also rarely encountered 
in a practical circuit, does indicate a maximum, or 
limit, cutoff current. 

The current which flows through a transistor when the 
base terminal is disconnected, and a current or voltage 
supply is connected across the emitter and collector 
terminals, will be forward current for one junction 
and reverse current for the other junction, depending 
on the polarity of the supply. When the polarity is 
such that the emitter-base junction is forward biased, 
the collector-base junction will be reverse biased. 
Reverse bias for the collector-base junction is the 
normal mode for transistor operation, and it is the 
correct conditio for measuring collector cutoff 
current and voltage breakdown with zero base current. 
The base terminal will be "floating" under these 


conditions, and there will be a floating voltage at 
the base terminal equal to the voltage-drop across the 
emitter-base junction. That voltage will increase 
with an increase in emitter current. 

cutoff The cutoff current which flows under these conditions 
current > is usually much greater than the reverse current 
reverse which flows through the collector-base junction when 
current the emitter lead is open (.Iq# ) • In fact, collector 
cutoff current with an open base terminal is as many 
times greater than collector-base reverse current 
(-^CBo) as tne value of tne forward current transfer 
ratio (for low values of collector current) . In 
other words, if beta is 50, then collector cutoff 
current with the base open will be 51 (50 + 1) times 
greater than the simple collector-base reverse current 
with emitter open. The reason for the current increase 
is that the reverse current through the collector- 
base junction has to be supplied from the emitter 
terminal, no other is available, and the carriers 
injected into the base region to supply that current 
diffuse and consequently allow many times that amount 
of current, to pass between emitter and collector. 

^(BR)CEO > Breakdown voltage for any given amount of cut-off 
7(RR)fB0 current with the base open is less than breakdown 
voltage for the collector-base junction alone 
(emitter open) . There are several factors that 
account for reverse current through a PN junction 
when a given reverse voltage is applied. All of 
which, except for surface leakage, are temperature 
dependent. When the surface leakage factor is 
negligible, reverse current will approximately 
double with every 6 C increase in temperature, for 
silicon transistors. For germanium transistors 
leakage will double about every 10 C. For this 
reason careful attention must be given to temperature 
when accurately measuring reverse current (cutoff 
current) . In some normal cases surface leakage 
current will dominate. 

I CE g and ^(br)ces — Collector Cutoff Current And 

Voltage Breakdown, Base Shorted 
To Emitter. 

The collector current, which flows when the base and 
emitter terminals are shorted together and reverse 
voltage applied, is a small fraction of that which 
flows when the base terminal is open. Under these 


test conditions most of the collector current passes 
through the base terminal rather than the emitter 
terminal because the base region in the transistor 
is adjacent to the collector region, and the external 
short offers less opposition to current flow than the 
internal emitter region. The collector cutoff current 
which flows is usually somewhat more than when the 
emitter terminal is open. The resistance between 
the material comprising the base region within the 
transistor and the base terminal causes a small 
voltage drop within the transistor that essentially 
forward biases the base-emitter junction a small 
amount. The forward bias permits the emitter to 
inject additional carriers into the base region and 
increase the total current. Measurement of collector 
cutoff current under these conditions is like 
determining what collector current will flow in a 
circuit when the transistor is driven from a very low 
impedance source, and the drive voltage is very close 
to zero. 

J CER and ^(br)cer — Collector Cutoff Current And 

Voltage Breakdown, Base Returned 
To Emitter Through A Specified 
Resistance . 

Jqcd falls When the base terminal is connected to the emitter 
between terminal through a resistor instead of remaining 
j anc j open or connected directly, the collector cutoff 

current will be some value between what it is when 
CES the base is open and what it is when the base is 

shorted to the emitter, at any given collector voltage. 
The resistor value may be selected to simulate the 
source impedance of a typical base-drive circuit to 
indicate what collector current would remain when 
the driving voltage went to zero. The collector 
breakdown voltage, ^br(CER) ' ^ or a gi ven cutoff 
current will be a voltage in between that for open 
base, Vbr(CEO). and that for shorted base, FbR(CES) • 

In-vvr and '^'(BR)CEV — Collector Cutoff Current And 

Voltage Breakdown, With 
Specified R&Ven'oei Voltage 

If a small reverse voltage is applied across the 
base-emitter terminals, collector cutoff current can 
Ir£\j < -^CES k e reduced below the value which flows with the base 
terminal shorted to the emitter terminal. 


This reverse base-emitter bias will also increase the 
voltage at which breakdown occurs, assuming of course, 
that breakdown voltage is measured at the same 
collector current value in both cases. Because some 
current may flow in the base circuit under these 
conditions, the resistance of the base circuit can 
cause the base terminal voltage to be less than the 
base supply voltage. Therefore, the base terminal 
voltage should either be measured at the base 
terminal or supplied from a very low resistance 
source. If the supply voltage and the resistance 
of the supply are known and specified the cutoff 
current could be classified as J CEX instead of I CEV . 

J CEX and 7 (BR)CEX "" Collector Cutoff Current And 

Voltage Breakdown, With 
Specified Base Drive Circuit. 

As has been shown by discussions of different 

conditions for the measurement of collector cutoff 

current, there may be a big difference in cutoff 

current depending on what, if any, external connections 

there may be between the base terminal and the emitter 

terminal. Predicting collector cutoff current in 

practical circuits may be simplified by using 

simulated circuits. Such measurements provide very 

good data for very similar circuits. The terms 

J CEX and ^BR(CEX) can be used instead of the terms 

^CEV and 7 BR(CEV) as lon S as equivalent conditions 
are stated. 

-^CEO greatest Collector cutoff current, although normally never 
with Vq high, great, may be different in a given transistor 
temperature depending on the conditions which may be said to 
high, and constitute cutoff. The highest amount of what could be 
IB zero be called cutoff current flows when a transistor 

has a high collector voltage, has zero base current, 
and is hot. The easiest way to assure zero base 
current is to leave the base terminal disconnected 
or open. Collector cutoff current for this condition 
is symbolically called J CE0 (0 for open base) . When 
the base terminal is not left open, but has a resistor 
of high resistance value connected externally between 
the base and emitter terminals, the base terminal is 
practically open but somewhat less cutoff current 
flows. As the resistance value of the resistor is 
reduced, cutoff current diminishes. Cutoff current 
measured under these conditions varies widely, 


depending on the value of the resistor, but is 
symbolized by the letters i CER (K for resistor in 
the base lead) . When the resistor value between the 
base and emitter is practically zero ohms, the cutoff 
current is represented by the symbol I"cES ^ S for snort 
circuit between base and emitter terminals) . 

Collector cutoff current can be diminished further by 
applying a reverse voltage between the base and 
emitter. The reverse voltage does not need to be 
great to do its job. One or two volts is usually 
adequate, when that voltage actually appears between 
the base and emitter terminals. The symbol for 
collector cutoff current for a reverse-biased emitter- 
base junction is I CEV (V for voltage between base and 
emitter) . The symbol 7"cEX could represent the same 
situation, as explained earlier. 

j CEV < j ces Very low cutoff current flows when a reverse voltage 
lr * l ' L is applied between the collector and base if the 

emitter is open. This configuration does not resemble 
any ordinary use of a transistor however, so it is 
somewhat of a misnomer to call it transistor cutoff 
current. It is really a measurement of a diode 
characteristic — the collector-base junction reverse 
current The symbol is I C bo (° f or °P en emitter) . 

7 CE0(SUS) 

Collector Sustaining Voltage 

A considerable error can be made in measuring cutoff 
current near the breakdown region of some transistors, 
because of a negative resistance characteristic present 
^CEX(SUS) in thls re 8 ion - In otner words, one of two distinct 
collector currents may flow with a given collector 
voltage applied, depending on how the collector 
voltage was chosen and applied, and whether the cutoff 
current was increased to the value selected or 
decreased to the value selected. With a transistor 
curve tracer both points can be shown and easily 
measuring distinguished. See Fig. 1-20 and Fig. 1-21. The 
errors curves in Fig. 1-21 changed from those shown in 
near Fig. 1-20 when the peak collector supply voltage was 
breakdown increased from 210 volts to 220 volts. With a 
region 220-volt supply, avalanche breakdown occurs with 

zero base current, and the collector voltage drops 
to 100 volts, but sustains about 180 mA with the 
particular load used. These curves show that when 
base current is switched between zero and 3.5 mA, 
collector current will switch between about 180 mA 


40 80 120 140 200 










o-r 1 








Fig. 1-20. Collector breakdown, 2N4111. 

at 100 volts, and 320 mA at close to zero volts, 
if the supply voltage is 220 volts. The average 
power dissipation of the transistor would be high. 
With a collector supply of 210 volts or less, the 
sustaining voltage would not be significant because 
the avalanche breakdown would not occur. In that 
case, the transistor would dissipate very little 
power because practically zero collector current 
will flow, even at 210 volts, when base current is 


COLLtClUK dKtAwjuwiM 








Tig. 1-21. 

Collector avalanche breakdown and 
sustaining voltage, 2N4111. 

neqative R 

From the shape of the zero base-current curve in 
Fig 1-21, different collector sustaining voltages 
and 'currents can be predicted for higher values of 
load resistance by using a corresponding load line. 

When there is a negative resistance characteristic in 
the breakdown region, it can be deLecled by slowly 
increasing collector voltage until the curve becomes 
vertical, then turns around and starts back. Peak 


current must be carefully limited with a high value 
series resistor to avoid the possibility of 
overheating the transistor. Under these conditions 
increasing the peak collector-supply voltage 
increases the peak collector current, which reduces 
the peak collector voltage. Any collector voltage 
In a (stable) negative resistance region sustains 
more collector current than when the peak collector 
voltage is first increased to that voltage under 
otherwise identical conditions. Similarly, when a 
transistor is switched from hard-on to hard-off, 
that is, operated with a high collector-supply voltage 
and high value series collector load resistor, the 
cutoff current will be relatively high, and the 
collector (sustaining) voltage relatively low. 

Sustaining voltage is the collector voltage required 
to sustain a given collector current where that 
current is the larger of two possible values at that 
collector voltage under a given set of collector-cutoff 
conditions. The principal purpose for identifying and 
measuring collector sustaining voltage is to indicate 
how the transistor may be operated without excess 
dissipation. Unless there is a negative resistance 
characteristic in the cutoff region there is no object 
in distinguishing between collector breakdown voltage 
and collector sustaining voltage. Most transistors 
exhibit no negative resistance characteristic for a 
cutoff condition where the base is open. But most 
do exhibit the characteristic when the base is 
reverse biased, shorted to the emitter, or connected 
to the emitter through any small value of resistance. 

Avalanche Breakdown 

avalanche: The term avalanche or avalanche breakdown is often used 
dynamic R synonomously with the term zener breakdown. Although 
drops both terms are carefully defined in the section of this 
book dealing with the definition of terms, the 
distinction needed for this discussion is between 
that breakdown where there is simply an abrupt 
reduction in the dynamic resistance of a device or 
junction, and that breakdown which is characterized 
by not only an abrupt reduction in dynamic resistance, 
but by a negative resistance characteristic. The 
measurement of the electrical characteristics of 
devices which have a negative resistance characteristic 
can sometimes be plagued with errors if the negative 
resistance characteristic is not known, or its 


existence not suspected. Even then control of the 
measurement may be difficult and uncertain. It is 
sometimes possible to plot graphically the complete 
resistance characteristics associated with collector- 
voltage breakdown using a transistor curve tracer. 
In cases where the negative resistance would be 
represented by a very radical change in slope, it 
may not be practical to swamp the negative resistance 
negative R by a comparable real resistance to plot a whole curve, 
is evident Nonetheless, avalanche breakdown does not go 
on curve- undetected on a curve tracer, and even when the 
tracer precise slope of the negative resistance region is 
not shown, the places where it begins and ends 
will be apparent. 

whenever a negative resistance characteristic is 
plotted on a curve tracer for a collector family, 
as when measuring collector sustaining voltage, 
for example, the negative resistance region will 
be unstable if the slope of the load line 
(established by the resistance in series with the 
supply) is too steep. As collector current is 
changed the load line should not intersect more 
than one point at a time on the curve in the 
negative resistance region. 

When not enough series resistance is used as collector 
current is increased into a negative resistance region, 
the current may suddenly step to a higher value, 
corresponding to lower collector voltage, at a point 
where the load line does intercept the negative 
resistance curve — after that curve has bent and 
become more steep. When such a step occurs, the 
transition is usually rapid enough to shock exite 
connecting leads into damped oscillations due to 
lead inductance and stray capacitance. Even if no 
oscillations occur there will be a section of the 
curve that has to be considered absent. That section 
will usually be easy to identify because it will be 
very dim -- indicating the CRT beam was deflected 
with exceptional velocity during the change from one 
current value to the next. 

Avalanche breakdown in some transistors can occur 
in extremely short intervals of time. When it can, 
then even small amounts of stray capacitance in 
the leads connecting the transistor will momentarily 
appear as a low impedance load to the transistor 
collector. In that case even though the current- 


limiting resistors in series with the collector supply 
may have a high enough value (to DC) to constitute a 
load line that would intercept the negative resistance 
curve at all points, high frequency parasitic 
oscillations may prevent plotting the whole negative 
resistance region. 

Some transistors are used for their very fast 
avalanche characteristics. See Fig. 1-22. 












Fig. 1-22. High-speed avalanche transistor. 
Selected 2N2501. 


40 80 120 160 200 

7 C£ (VOLTS) 

Fig. 1-23. Breakdown due to reach- through, 20yA 
base current steps, 2N3877A. 

Punch-Through Or Reach-Through 

The terms punch-through or reach-through apply to a 
collector-breakdown condition where base current has 
little or no influence on the collector-breakdown 
voltage. The condition only applies to some 
transistors, usually ones with a very thin base 
region. With transistors of this type the collector- 
base junction depletion region may extend all the 
way through the base material into the emitter 
matprial, when enough collector-to-emitter voltage 
is applied; before some other form of breakdown 
occurs. When this happens a good conduction path 


between collector and emitter Is created at a 
particular collector voltage, directly through the 
base material. Increasing collector voltage beyond 
that value causes a radical increase in collector 

punch- Breakdown due to the collector-voltage field 
through extending through the base material is easily 
diagnosis recognized on a transistor curve tracer. Whenever 
the breakdown current is the same value regardless 
of the base current, a transistor curve tracer will 
show different curves in a collector family joining 
at the breakdown region. Fig. 1-23 displays peak 
collector sweep voltage limited enough to show only 
the bottom two curves joining. However, all five 
curves merge when collector breakdown current was 
allowed to increase enough. 

Reach-through is a non-destructive form of breakdown 
as long as collector current is limited. 


The speed and fidelity with which a transistor may 
fully respond to a sudden discrete change in input 
current has a relationship to its gain characteristics 
at high frequencies — but not a simple relationship. 
While it is generally true that the transistors which 
have the better high frequency characteristics are 
also the ones which are faster in their response to 
step signals, the best transistors for high frequency 
sinewave amplifiers don't always make the best 
switching circuits. This is particularly true 
when the kind of switching to be done involves 
driving the transistor into saturation or into cutoff. 
When the drive is alternately between saturation 
and cutoff, the correlation is at its worst. 

switching- Even if and when all of the several individual 
transistors: characteristics about a transistor that determine 
measure its step response are quantitatively known, 
vs predicting the performance of the transistor by 

calculate use of mathematical equations is laborious and 

approximate. Testing and measuring step response 
or switching time of transistors is therefore very 


Turn-On Tvne Ana inrn-ujj i-uhv 

The semiconductor industry has adopted several terms 
related to the switching time of transistors. First 
of all, a distinction is made between the time it 
takes a transistor to turn on and the time it takes a 
turn-on transistor to turn off. One is called turn-on time 
= delay plus and the other called turn-off time. Both the turn-on 
risetirne time and the turn-off time are divided into two 

intervals, each described by a separate term. Turn-on 
turn-off time is divided into delay time and risetirne. 
= storage Turn-off time is divided into (carrier) storage time 
p| us and falltime. One should remember from the outset 
falltime that here risetirne and falltime have nothing to do 

with whether the waveform being measured is positive- 
risetime going or negative-going. Risetirne applies to an 

- Jp increase increasing collector current. This time will be 

. coincident with a negative-going collector voltage 
falltime waV eform for an NPN transistor, or a positive-going 

- I c decrease coiiector vo ltage waveform for a PNP transistor. 

Falltime applies to decreasing collector current. 

tjj — delay Time 

Delay time is the time between the instant when a 
current step is applied to turn the transistor on, 
and the instant when collector current has increased 
define delay to 10% of its final value. To avoid any ambiguity 
time and about the instant when a current step is applied, 
carrier measurement of delay time should be made starting at 
storage time a point on the applied step 10% of the way to its 
final level. 

Carrier storage time is the time between the instant 
when base current is cut off, and the instant when 
collector current diminishes to 90% of its full value. 
Again, to make the definition and measurement of 
carrier storage time more precise, the instant when 
base current is cut off is said to be when it has been 
reduced to 90% of its full value. See Fig. 1-24. 

From this figure showing idealized wave shapes one may 
erroneously infer that the falltime of the output 
pulse is about the same as the 90% to 10% falltime 
of the applied pulse. One may even infer that 
risetirne and fallrfme are usually about the same. 
No implications of this kind were intended. It is 
typical, however, for the risetirne and falltime of 









Fig. 1-24. Switching transistor pulse 
characteristic . 

the applied pulse to be close to the same. It is also 
appropriate that the risetime and falltime of the 
applied pulse be much shorter than the response 
stimulated. Otherwise it would not be possible to 
discern the true response limitations. 

Whenever delay time is a significant portion of 
turn-on time it is principally due to one or a 
combination of two factors, both associated with the 
quiescent cutoff condition. 

Whenever the emitter-base junction is reverse biased 
to create the cutoff condition, the emitter-base 
junction capacitance will be charged to the 
reverse bias voltage, and constitute a charge that 
will have to be removed before conduction can start. 
Of course, the same is true for the stray capacitance 
of the emitter and base leads, both inside and outside 





12ns WIDE, 


40 80 120 


Fig. 1-25. Turn-on characteristics of small 
high-speed transistor. Six 
collector voltage curves that 
correspond to six base bias levels 
between cutoff and saturation show 
various rise rates. Turn-on delay 
(£j) apparent for only top curve; 
collector load resistance 2S0fl as 
in Fig. 1-28. Risetime corresponds 
to falling portions of curves. 


of the transistor package. But this capacitance is 
only of significance when the transistor is back- 
biased into cutoff. If cutoff is not so hard — as 
when the base current or base voltage is only reduced 
near zero, there is only one principal factor. That 
is diffusion time — the time it takes carriers 
injected into the base region to diffuse and let the 
base region become a good conduction medium for 
collector current. A transistor which is switched 
out of cutoff, but not into saturation, will exhibit 
as much delay time as if it were switched into 
saturation. Delay time is usually minimal when the 
transistor is barely cutoff, and then is turned on 
hard. In Fig. 1-25 we have shown the collector 
waveforms that result when a fast-rise turn-on pulse 
approximately 12 ns wide and 2 volts tall is applied 
to the base of an NPN transistor through a 
non-inductive 1000-ohm resistor. An oscilloscope 
rather than a curve tracer was used. The transistor 
was biased at six different levels including 
saturation and cutoff. Only the top curve represents 
a quiescent cutoff condition. The bottom curve shows 
the applied pulse and the curve next to the bottom 
represents the transistor biased into saturation 
before the turn-on pulse was applied. Notice that 
none of the curves show any turn-on delay except the 
top curve, where about 1 minor division (or 4 
nanoseconds) of delay is apparent. 

t T — Risetime 

Risetime (the negative slope on the collector 
waveforms in Fig. 1-25), cannot be measured from 
these curves because the turn-on pulse did not last 
long enough. But the curves illustrate other things. 
Even though each curve represents the same increase 
in base current (2 mA) a considerable difference in 
slope and amplitude can be seen for the different 
curves, particularly as collector voltage approaches 
saturation. This should show us that the change in 
slope is due to a change in collector voltage. The 
top curves show a relatively linear rise rate 
because collector voltage saturation has not been 
reached. When these curves are allowed to continue 
toward saturation by making the turn-off pulse last 
longer they also become rounded, but it is difficult 
to tell from those curves alone whether the rounding 
reveals a basic RC type of characteristic or not. 
Risetime is primarily limited by the collector-base 
junction capacitance, and all stray capacitance 
between the collector and base leads, plus the 


amount 01 oase tuiieai avdiiauic w uiscnaigc "'i^ 
capacitance. When the collector voltage changes as 
a result of changing the collector current — as it 
will unless the load impedance is zero — the 
collector-base capacitance is, in effect, increased 
in proportion to the change in collector voltage. 
input The magnitude of the increase is limited by the 
capacitance reverse voltage transfer characteristics of the 
change transistor, and is approximately equal to the ratio 
A7 r of the change in collector voltage to the change in 
base voltage. This figure can be very large. In 
essence, current in the base lead must discharge the 
collector-base capacitance as the collector voltage 
decreases. So, when current in the base lead is 
suddenly increased to a new level, most of the 
increase is at first diverted to discharge the 
collector-base capacitance. This delays the 
increase of carriers in the base region, which 
accounts for most of the risetime. The rise rate 
is linear except in the region near saturation, 
because a practically constant current discharges a 
practically constant capacitance. Collector-base 
junction capacitance will typically increase as the 
collector voltage reduces, however, and this also 
accounts for a slope that is less steep for lower 
values of collector voltage. 


The importance of collector-base junction capacitance 
virtual as a limiting factor on transistor risetime was 
input discussed. The fact is emphasized that the actual 
q » values may be only a small fraction of 
actua I C tlle virtual capacitance which must be discharged by 
base current. Stray capacitance, due to circuit or 
socket capacitance, between collector leads and base 
leads can account for a poor correlation of 
measurements between otherwise identical test 
fixtures when measuring risetime or rise rate. 

Some people will prefer measuring rise rate to 

measuring risetime when evaluating the turn-on time 
dV of a transistor. Rise rate would be expressed in 

cit terms of volts per unit time, but might be measured 

by determining the time it takes for the slope to 

cross two discrete voltage levels. 

t — Carrier Storage lime 

Fig. 1-26 is similar to Fig. 1-25 except that a 
turn off pulse was applied instead of a turn-on 
pulse. The applied turn-off pulse is shown on the 
bottom curve to establish when turn-off commences. 






12ns WIDE, 


40 80 120 



Fig. 1-26. Turn-off characteristics of small 
high-speed transistor. Five 
collector-voltage curves that 
correspond to five base bias levels 
between saturation and cutoff. 
Carrier storage time (t s ) apparent 
in curve 5 is partly due to 
quiescent saturated condition. 
Conditions similar to Fig. 1-25 and 
Fig. 1-28. Turn-off pulse is 2mA 
(Two volts across 1 kn) . 


1~L1L. VtO Cli-t, WX. l-liv- oUniL. MJ.H w j-i-mu^w ^- w i 

as used for Fig. 1-25 and the beginning of turn-off 
is represented by the up-going portions of those 
curves. The curve adjacent to the bottom curve is 
the only curve representing a quiescent saturated 
condition. Notice that this curve is the only one 
which does not appear to start to respond the 
instant the turn-off pulse is applied. Because the 
turn-off pulse is only about 1.2 volts in amplitude, 
and it can reduce base current by only about 1.2 iA 
(through a 1 kfi series resistance) , the collector 
could not be saturated very much and still show the 
influence of only a 1.2 mA reduction in base current. 

See Fig. 1-27. To show a delay in response to a 
turn-off pulse, the drive pulse amplitude was 
increased so that the transistor could be saturated 
harder and the turn-off pulse still be able to reduce 
base current enough to let the transistor come well 
out of saturation. The time scale was reduced from 
20 nanoseconds per division to 5 nanoseconds per 
division, to avoid crowding the curves. The four 
top curves are produced under identical conditions 
except that the quiescent bias current was changed. 
The top two curves represent bias levels that allowed 
the transistor to remain out of saturation. These 
two curves show essentially no delay in response to 
the turn-off pulse. However, the bottom two collector 
curves represent a saturation condition, and show a 
delay in response to the turn-off pulse. The curve 
which shows the greater delay represents the more 
£ s ~ storage saturated condition. This delay in response to 
ttme turn-off is called carrier storage time, or simply 
storage time (t ) . As the term implies, the delay 
is attributable S to an excess of carriers someplace. 
The excess carriers are primarily in the base 
material unless the collector-base junction becomes 
forward-biased in saturation. The primary reason for 
the excess is that collector voltage has been reduced 
so greatly by the voltage drop across the collector 
load resistor, that an insufficient voltage remains 
to collect all of the carriers that have become 
mobilized. These excess carriers will eventually 
disappear due to collector current, and by 
recombination with others of opposite polarity within 
the semiconductor material. Storage timp will be 
less if new carriers are not created by allowing some 




10 20 30 



Fig. 1-27. Turn-off characteristics of small 
high-speed transistor using 6mA 
turn-off pulse. Four collector-voltage 
curves corresponding to four bias 
levels. Turn-off corresponds to 
rising portions of curves. Curves 
C and D show storage time (t s ) , a 
delay in response to turn-off pulse, 
due to beginning in saturated 
condition. See Fig. 1-28 for circuit. 


cutoff pulse residual base current to continue to flow. In other 
hiyii and words, the applied cutoff pulse will reduce carrier 
fast reduces storage time if it is high in amplitude as well as 
t fast in transition time. When the turn-off pulse is 

b more than tall enough to reduce base current to zero, 

the direction of current flow in the base lead will 
reverse and tend to back-bias the emitter-base 
junction. This momentary reverse current will speed 
up removal of excess carriers, too. 

It is worthwhile noting that the carrier storage time 
represented by the middle two curves in Fig. 1-27 
far exceeds the risetime of the leading edge of the 
turn-off pulse. If the risetime of the turn-off 
pulse had not been much shorter than the storage 
time, there could be a considerable difference in 
the display and the measurement of storage time. 

t — Fall Time 

fall time: Fall time is determined by essentially the same 
same factors factors that determine risetime. Refer to the 
as risetime paragraphs on risetime for that discussion. Figs. 
1-25 and 1-26 show the same transistor under nearly 
identical test conditions, except Fig. 1-25 shows 
response to a turn-on pulse whereas Fig. 1-26 shows 
response to a turn-off pulse. Compare the falling 
slopes (risetime) of Fig. 1-25 with rising slopes 
(fall time) of Fig. 1-26. 

Fig. 1-28 shows a simplified diagram of the test 
circuit used to produce the curves photographed and 
shown in Figs. 1-25, 1-26 and 1-27. 


WITH 5Q0) 


950 ,_ OUTPUT 

Fig. 1-28. Simplified switching- time test 



les lbs leo lbo 

The input capacitance of a transistor will depend on 
ies - input, whether the transistor is operated in the common- 
common- emitter mode or the common-base mode, and it will 
emitter, zero depend on the value of the load impedance. The 
U output symbols C^ es and Cj^s are used when the output load 
i bs - input, is zero ohms (output short circuited). To symbolize 
common-base input capacitance under the opposite extreme of 
zero Q output output load resistance (output open circuited), C-£ eo 

and C ibo are used. The symbols, therefore, stand 

for boundary conditions that cannot quite be 

realized in practice in any case. 

The input capacitance of amplifying devices like 
vacuum tubes and transistors is of interest to a 
circuit man because that capacitance has a bearing 
on the high frequency response or switching time of 
his circuit. In particular it will give a clue to 
the loading effects on whatever is used to drive the 
input of the amplifying device. With vacuum tubes, 
which are voltage-controlled devices ideally having 
a high input impedance under all conditions, input 
capacitance will predominate at even audio 
frequencies when the driving impedance is high. 
With transistors, that are current-controlled and 
have an input impedance that is low compared to 
vacuum tubes, input capacitance has a somewhat 
different significance. 

To a very limited extent the plate-grid capacitance 
of a triode is comparable to the collector-base 
transistor junction capacitance of a transistor. And to the 
vs triode same limited extent the grid-cathode capacitance is 
comparable to the emitter-base junction capacitance 
of a transistor. These two capacitances plus 
voltage gain are practically all that need be known 
about a vacuum tube to predict its input capacitance. 
With transistors, however, several things are 
radically different: 

1. Collector-base junction capacitance changes with 

collector-base voltage, 
real C and 2. Emitter-base junction capacitance may have very 
virtual C little change in charge because base-emitter 

voltage typically changes slightly when the 

transistor is turned on. 


3. Carrier diffusion time acts like capacitance. 

4. Carrier recombination time acts like capacitance. 

5. Some of the real and the virtual capacitances 
have considerable resistance to charge and 
discharge through. 

Because of the equal importance of these other various 
characteristics, input capacitance is usually 
estimated by measuring the two junction capacitances 
using a capacitance bridge. Input impedance or 
admittance measurements at various frequencies under 
various operating conditions using RF bridges are 
used fcr a more complete picture. Bridges are 
offered commercially that permit high frequency 
measurements to be made beyond 1 GHz. 


C , C , , C and C, 
oes obs oeo ODO 

The output capacitance of transistors, similar to 
output C vs input capacitance, depends on a complex variety of 
HF things. Output capacitance is also of interest 

performance primarily because it affects high frequency 

performance. Besides measuring junction capacitance 
with a bridge, output admittance measurements are 
sometimes made at various frequencies from which most 
high-frequency-limiting factors may be deduced. 
Admittance and impedance bridges are offered for these 
kinds of measurements that perform well beyond 1 GHz. 


As with the measurement of input and output 
admittance at high frequencies, measurement of 
transfer functions at high frequencies can be made 
with bridges designed for that purpose. 

f — nut-Off Frequency for h 

The small-signal current transfer ratio for 
transistors operated in the common-emitter mode 
invariably becomes smaller and smaller at higher 
and higher frequencies. The frequency at which the 


current gain of a transistor driving a low impedance 
load decreases by 3 decibels (current down 29.3%) is 
the cut-off frequency. For example, a transistor 
having a current gain of 100 at low frequencies would 
have a current gain very close to 70 at the cut-off 
frequency. There would still be a very considerable 
gain at the so called "cut-off" frequency. 

/ T — Frequency of Unity Current Gain, Common-Emitter 

As the small-signal short-circuit current gain of a 
transistor is measured for signals having higher 
frequencies than f^f e , a frequency can be found where 
the gain has diminished to unity, or one. At higher 
frequencies than that the current gain is less than 
/ T - no gain one, that is, a loss. So / T can be considered the 
frequency beyond which the transistor will not 
provide current gain. 

The frequency fj can usually be determined using 
equipment that does not extend to / T in direct 
frequency measurement capabilities. The reason is 
that current gain usually falls off at the rate of 6 
db (50%) every time the frequency is doubled, beyond 
the cut-off frequency. The roll-off curve is usually 
very similar to a simple RC curve. From such a curve 
fj may be extrapolated. Usually a checkpoint or two 
is selected at one or two frequencies beyond the 
cut-off frequency to assure a more reliable 
9 ain calculation. The symbol fj is sometimes called the 
bandwidth gain-bandwidth-product frequency, because the 
product product of gain and frequency will be a constant 

for frequencies beyond /x and fairly constant for an 
octave or two below _f T when hf e is high at low 

/ hf b — Common-Base Cut-Off Frequency 

The common-base mode does not provide a current gain. 
Gain is typically very close to one but never is 
more than one. Nonetheless, even this low gain 
falls off at high frequencies. The frequency at 
which the gain falls 3 db below that for low 
frequencies and DC is fhfb- This cut-off frequency 
is usually in the same vicinity as fj. 



Bipolar transistors are current-controlled 
semiconductor devices, and have low input resistance 
compared to vacuum tubes and field-effect transistors. 
bipolar To calculate the loading effect that a transistor 
transistors will present to whatever driving source it may be 
have low connected to requires a knowledge of its input 
Z input characteristics. Except at high frequencies where 

input capacitance may influence input impedance, and 
except for extremely high frequencies where 
transistor lead inductance may enter the picture, the 
input characteristics of a bipolar transistor are 
Z input - essentially resistive in nature. Measurement of h ±& , 
h. h. h. h ±h , and h — the small-signal h-parameter common- 

' e | b ' c emitter, common-base, and common-collector symbols 
- R input for ±nput impedance — pertain strictly to resistance, 
because they apply at only low frequencies and DC. 
Input resistance is, however, typically nonlinear if 
measured over a wide range of driving current. 

r input It is because of the nonlinear nature of input 
nonl inear resistance that we must be specific about the 

conditions for measuring it. Any transistor can be 
operated in a variety of ways to have a wide range 
of input resistance values. The common-collector 
mode offers the highest input impedance by quite a 
margin, extending into megohms, when the load is also 
a high impedance. The common-base and common-emitter 
modes have low input impedance values. This range may 
be from a fraction of an ohm to over 1000 ohms 
depending on the amount of forward bias, and whether 
the collector voltage is near saturation. The 
common-base configuration has the lowest input 
impedance by a considerable margin, for comparable 
amounts of collector current. 

fc IE and h ±e — Static and Dynamic Input Resistance, 

Static input resistance, h^, is equal to the quotient 
V of the voltage-drop across the base-emitter terminals 

h = _BE and the base current producing that voltage drop — at 
IE J B a given collector voltage. It can be easily 

calculated after the current and voltage have been 
measured with simple DC instruments. Or the currents 
may be applied and voltages read from the curves 
presented by a transistor curve tracer. See Fig. 1-29. 







Fig. 1-29. Input resistance, 2N1304. Five 

constant current base steps applied. 
Voltage between base and emitter 
terminal 20 mV/div -- offset 120 mV. 
Dynamic input resistance (hie) 
between point A and point B = 20 mV 
(AV BE ) divided by .05 mA (A!"b) which 
equals 400fi. 



From the same display one may also determine dynamic 
(differential) input resistance, h i& . The symbol h ±e 
stands for dynamic input resistance measured with 
the output load essentially zero ohms. The term 
"small-signal short-circuit" implies that the 
collector voltage should remain constant when the base 
current is changed. When measured on a curve tracer 
therefore, a vertical line at any horizontal position 
corresponds to one particular collector voltage. The 
vertical distance between curves at the horizontal 
position corresponding to a given collector voltage 
is proportional to the change in voltage drop at the 

base induced by a selected change in base current. 

AJ Dividing the measured change in base voltage, AF BE , 
R by the selected change in base current, AJ B , gives 
the differential input resistance for that set of 

By looking at the set of curves and noting the 
difference in vertical distance between each of them 
you can instantly perceive the difference in input 
impedance at various bias conditions. The vertical 
distance between adjacent curves is directly 
proportional to the dynamic input resistance at any 
given collector voltage. 

h and h ,_ — Static and Dynamic Input Resistance, 

IB lb 

* " BE 

Static input resistance, h 1B , is equal to the 
-- — quotient of the voltage drop across the emitter-base 
IB I £ terminals and the emitter current. The collector 

voltage will have a small influence so it should be 

The voltage-drop across the emitter-base terminals is 
due to base current, the same as for the common- 
emitter mode, not due to emitter current. Therefore, 
if we were to start with the same amount of base 
current as we did for a measurement of input 
impedance in the common-emitter mode, we can compare 
input impedances of two modes. Under these 
conditions we would compare the quotient of V^/1% 
measure^ with the quotient of 7 B e/J"e- This is like simply 
and hff, comparing base current (I B ) with emitter current (I E ) 

calculate In other words, input resistance for the common-base 
h mode is much less than for the common-emitter mode, 

when other conditions are the same. 


h IE 

IB ft FE + 1 

So ?z IB may be determined by measuring ft-j-g and 7z FE , 
then calculating ?Zt B . 

^■[■q and hj_ c — Static and Dynamic Input Resistance, 

The input impedance for the common-collector 
configuration is higher than the common-emitter 
configuration, except when the load is zero ohms. 
When the load is zero ohms the input resistance is 
the same as for the common-emitter mode. Therefore, 
hjQ and h^ c , which are symbols for static and 
dynamic input resistance with output short-circuited, 
can be measured using the common-emitter configuration. 

Input Impedance of Emitter-Followers 

The load, in the common-collector mode, is driven by 
the emitter rather than the collector. The circuit 
configuration is frequently called an emitter- 
follower, unless the load is close to zero ohms. 
Emitter-followers are seldom intended to drive zero 
ohms, so in practice, the input resistance is higher 
emitter than for the common-emitter mode. The voltage at 
output the emitter terminal, because it must remain close 
fol lows base to the voltage at the base terminal except when the 
input transistor is cut off, will move up and down closely 
following the voltage excursions on the base. As 
long as the voltage excursions don't go beyond the 
operating voltage level to which the emitter load is returned, 
limits where cutoff or breakdown would occur, or too close 
to the level of the collector supply voltage, where 
saturation would occur, an emitter follower may be 
made to operate normally. 

When the load driven by an emitter follower is high 
compared to the input resistance of the same 
transistor operated in the common-emitter mode, the 
input resistance is approximately equal to the 
product of the resistance of the load (i? L ) and the 
common-emitter forward current transfer ratio (hr,J) . 

The static value of input resistance differs somewhat 
from the simple product of h^ and i? L because the 
collector-to-emitter voltage does not remain 


constant as collector current changes in an emitter- 
follower. When current increases in an emitter- 
follower, the emitter voltage approaches the 
collector voltage and thereby reduces the current 
transfer. Likewise when base current is reduced and 
collector current is also thereby reduced, the 
collector-to-emitter voltage increases. The net 
increase in collector voltage retards the reduction 
in emitter current. 

The combined effect can readily be discerned and 
measured on a transistor curve tracer by connecting 
the selected value of load resistance between the 
emitter terminal and ground then displaying a family 
of curves depicting collector current versus collector 
voltage. See Fig. 1-30. At a collector-to-ground 
voltage of 5 volts, 4 mA of collector current flc 
20 uA/s+ep when base current is 0.12 mA (6 x .02 mA) . The 
x 6 steps static input resistance at this point (Point A) c 

the curve would be equal to the base-to-ground voltage 
divided by the base current. The base current 
(0.12 mA) plus the collector current (4 mA) flows 
through i? L producing 4.12 volts of drop. So the base 
voltage will be close to 4.12 volts. Actually it will 
be about 0.6 volts more than that because the base- 
emitter junction is forward biased. The static input 
impedance is, therefore: 



4.72 volts 
,00012 amperes 

40,000 ohms 

This operating point is very close to saturation, as 
can be seen by the knee on the curve just below the 
5-volt collector voltage point on the curve. This 
transistor would not normally be operated as an 
emitter follower with more than 4 volts on the base 
unless it were operated with a higher collector 
voltage than 5 volts. 

The differential input resistance would depend on how 
the transistor was biased in the quiescent condition 
and how big the changes of input voltage are. If we 
consider the quiescent condition to be with .06 mA of 
base current and 5 volts on the collector, the mid- 
section of the middle curve (Point C) in Fig. 1-30 is 
representative. Collector current is 1.7 mA and base 
current .06 mA, so emitter current is I. lb mA. This 
means an emitter voltage of 1.76 volts (voltage drop 
across 1 kilohm) and a base voltage very close to 0.6 
volts higher than that, or 2.36 volts. If we raise 


A B C 


Fig. 1-30. 2N3137 emitter follower conductance 
characteristics. R L equals 1000ft. 
Base current steps are 20 uA. 

input R 



the base-to-ground voltage enough to increase base 
current from .06 mA to .08 mA (Point B) , collector 
current will increase from 1.7 mA to 2.4 mA, a change 
of 0.7 mA. Emitter current will change from 1.76 mA 
to 2.48 mA, a change of 0.72 mA. If we assume that 
the base-emitter forward voltage drop remains 
essentially constant, which it does except near 
cutoff, the change in emitter voltage is a good 
measure of the change in base terminal voltage. The 
change in emitter voltage is proportional to the 
change in emitter current and is equal to the product 
of the load resistor, 1 kilohm, and the emitter 
current change, 0.72 mA. This product equals 0.72 
volts. The differential input resistance is equal to 
a change in base-to-ground voltage divided by the 
accompanying change in base current. By applying a 
known base current increase and calculating the base 
voltage increase, the differential input resistance 
is determined. In the example the dynamic input 
resistance is: 

0.72 volts 
.00002 amperes 

36,000 ohms 


The same measurement can De peiioimeu wnen T-suucvrtg 
Input voltage and current as when increasing input 
voltage and current. The vertical separation between 
the curves at a particular collector-to-ground 
voltage is proportional to the differential input 


Admittance is the reciprocal of impedance so 
measurements of output admittance are similar to 
measurements of output impedance. A high impedance 
is the same as a low admittance. Impedance is 
equal to voltage divided by current so admittance 
is equal to current divided by voltage. 

/i — Dyriamic Output Admittance, Common-Emitter 

The output admittance, or output impedance, of a 
transistor tells us what effect on current through 
the transistor a change in the output terminal 
voltage may cause. Measurement of a transistor's 
output admittance is usually done by simulating a 
constant-current or "open-circuit" input. 
Transistor curve tracers simulate this condition 
very well. 

Any curve which is a plot of current versus voltage 
has a slope which is equivalent to some impedance 
and some corresponding admittance. The collector 
current versus collector voltage curves that 
represent different discrete values of base current 
have two principal slopes. The slopes below the 
knees represent saturation resistance . Above the 
slope above knees of the curves, the slopes represent output 
knees for admittance or output impedance. The more nearly 
Z or Y horizontal a section of a curve is, the less will be 
out out t ^ e output admittance it represents, provided 

current is plotted on the vertical axis and voltage 
is plotted on the horizontal axis. Most curves on 
most transistor curve tracers depict current on the 
vertical axis. 

Small-signal output admittance for the common- 
emitter mode is measured at a given base current and 
between two given cullecLoi voltages. The difference 
in collector current that would accompany the change 


in collector voltage represented by the difference 
between two given collector voltages is proportional 
to output admittance. For the common emitter mode: 

AJ c 

^oe ~ Tv~ when base current is kept constant. 

See Fig. 1-31. 







Fig. 1-31. Output admittance (h oe ) depends on 
slope of curves. /\Vq and Mq 
between point A and point B determine 
output admittance for base current of 

250 yA. 


When the slope of a curve is nearly horizontal, 
expanding the display vertically so only a portion 
remains on screen will accentuate the slope and 
permit more accurate measurements to be made. See 
Fig. 1-32. 



.4 .8 1.2 1.6 2.0 

7 CE (VOLTS) ► 



Fig. 1-32. Output admittance measurement 

accuracy improved with 10X vertical 
magnification. i\Iq = .18 niA; 
AV rE = 1 V. 




Dynamic Output Admittance, Common-Base 

The measurement of the output admittance of a 
transistor operated in the common-base configuration 
is done in very much the same way as for the common- 
emitter configuration. The common-base mode has a 
lower output admittance than the common-collector mode 
or the common-emitter mode. It follows that the 
output impedance for the common-base mode is the 
highest of the three modes. The slope of the curves 
which represent graphs of collector current versus 
collector voltage for various discrete amounts of 
emitter current is very nearly horizontal, as can be 
seen in Fig. 1-33. Compare Fig. 1-33 with Fig. 1-34. 



Fig. 1-33. Output admittance (h ^) depends on 
slope of curves. See Fig. 1-34. 




• 10 12 3 4 5 

V m (VOLTS) 




Fig. 1-34. Output admittance (,h ^) . AJ C 
.08 mA; AF CB = 3 V. 


Output admittance is the quotient of a collector 
voltage change and an accompanying collector current 
change at some region out of saturation. Usually 
emitter current is held constant. For the common-base 

ob AV when emitter current is kept constant. 

^ oc — Dynamic Output Admittance, Common-Collector 

The measurement of h QC is made by measuring h oe for 
equivalent conditions and considering the two equal. 


Reverse-voltage transfer is a little like forward- 
current transfer; each is expressed as a ratio of a 
change at an output terminal compared to an 
accompanying change at an input terminal. Unlike 
forward current transfer, where the change in the 
output current is naturally considered a result of 
a change in the input current, reverse voltage 
watch it! transfer is the opposite. A change in output 

voltage can be considered to cause a change in input 
voltage when input current is held nearly constant. 
The reverse voltage transfer characteristic is 
important because a change in current through a 
transistor does often cause a considerable change 
in voltage at the output terminal, depending on the 
load, and this voltage change may affect the 
operation of the transistor. 

The measurement of the small signal reverse voltage 

transfer parameters 7i_ - h , and h requires the 

re rb re 
input current to be constant. This condition is 

comparable to the current being supplied through a 

very high resistance — ideally an infinite resistance 

or "open-circuit." 

& re — Reverse Voltage Transfer, Common-Emitter 

Measurement of the reverse voltage transfer ratio 
for the common-emitter mode using a transistor curve 
tracer may be accomplished by plotting base voltage 
versus collector voltage for various discrete base 
currents. The slope of the curves any place beyond 
the knees will be an indication of the reverse 
voltage transfer ratio. The measurement is made by 





Fig. 1-35. Reverse voltage transfer (h Te ) 

2N1304, at base current of .25 mA. 
A^ce between point A and point B is 
4 volts. A7gE between same points is 

approximately 4 mV. h re - " 




first finding the appropriate points on the 
appropriate curve that constitute the difference in 
collector voltage required, then measuring the 
difference in base-emitter voltage drop that 
corresponds to those points on the curve. Dividing 
the collector voltage difference (A7 C ) by the base 
voltage difference (A7 BE ) yields the transfer ratio. 
See Fig. 1-35. 

\b "" Reverse Voltage Transfer, Common-Base 

The measurement h rh can be made in a way similar to 
measuring h re , but the accuracy will be even less 
because the curves will be more nearly horizontal. 

\ c — Reverse Voltage Transfer, Common-Collector 

The reverse voltage transfer ratio for the common- 
collector mode is typically very close to one (1) 
because h r& is typically a very small fraction. 

h = 1 - h 
re re 




I and F, , — Leakage Current and Breakdown 

The measurement of leakage current and breakdown 
voltage of field-effect transistors is usually made 
with voltage applied between the gate and the source 
or between the gate and the drain. Or the drain 
may be shorted to the source, and voltage applied 
between the gate and the common connection between 
drain and source. The symbol ^(br)gSS stan ds for 
gate-source voltage breakdown measured between the gate and the 
breakdown source with the remaining terminal (the drain) 

shorted to the source. Measurement of breakdown 
voltage of insulated-gate field-effect transistors 
should ordinarily not be attempted. When breakdown 
does occur on this type of field-effect transistor, 
damage to the transistor may ensue even when 
destructive current is limited. They may, however, be tested 
testing to see that breakdown does not take place below a 
certain rated voltage. 





(mA) -04 



20 30 



. - 



— wv 





Fig. 2-1. ^ (br) GSS 2N4416 junction FET. 


Fig. 2-1 shows the breakdown region of an N-channel 
junction field-effect transistor displayed using a 
transistor curve tracer. From the photograph, 
breakdown can be seen to be at about 43 volts. Less 
than one volt increase in that region causes a current 
increase from less than 1 microampere to more than 
100 microamperes. The manufacturer says the 
breakdown voltage should be no less than 30 volts 
at 25 C (room temperature) . Breakdown is considered 
to have occurred for this transistor when 1 
microampere of current or more flows. 

Leakage current is measured between the gate and 
source also, usually under the same set of conditions 
as for measuring breakdown voltage, except somewhat 
less reverse voltage is applied. The symbol I G gg 
stands for leakage measured under these conditions. 

Leakage current measurements may call for measurement 
of picoamperes (10~^2 amperes) . DC Meters are 
leakage usually required for measurement of currents in the 
measured in picoampere range. The desired, or specified, reverse 
picoamps voltage is applied, and the meter connected in series 
to read the current. Some transistor curve tracers 
will measure leakage current down to somewhat less 
than 1 nanoampere (1 x 10"^ amperes) . 

Drain Breakdown 

Field effect transistors, like bi-polar transistors, 
have another kind of breakdown mode, one which occurs 
during the normal kind of operating conditions, with 
drain-to-source voltage applied and drain current 
flowing. Under these conditions breakdown starts 
when drain current increases greatly as a result of 
only a small increase in drain voltage. The symbol 
^(BR)DSS is used t0 indicate drain breakdown voltage 
when the gate-to-source voltage is zero, gate 
shorted to source. 


"gs = 0V 



7 (BR)GSS 


Fig. 2-2. Drain breakdown 2N4416 junction FET. 

Fig. 2-2 shows breakdown starting somewhere between 
about 42 volts and 45 volts. The same junction 
field effect transistor was used for Fig. 2-2 as for 
Fig. 2-1, where breakdown was about 43 volts. This 
is not merely coincidental. Whenever the draiu-to- 
source voltage exceeds the breakdown potential, the 
input current will add to the output current. 
Temperature affects the breakdown voltage somewhat. 
The breakdown voltage increases when temperature 
increases. This can be shown by comparing Fig. 2-3 
wiLl. Fig. 2-2. In Fig. 2-3, because the input 
voltage (Vq§) is always zero volts, the average 
drain current is much higher than when the input 


J increase 



voltage reduces drain current by more than half that 
value for more than half the time. The higher average 
drain current in Fig. 2-3 causes the field effect 
transistor to be hotter. Incidentally, increased 
temperature causes a reduction in drain current — 
other conditions being equal. Notice in Fig. 2-3 
that zero bias drain current is less than 6 mA when 
drain voltage is 10 volts, and more than 6 mA under 
the same conditions except for temperature — shown 
in Fig. 2-2. The loops on the high-current curves 
are an indication of the thermal time-constant for 
the field effect transistor. Temperature increases 
as drain current increases, as a sweep voltage 
increases. That tells us the field effect transistor 
is hotter when the sweep voltage is reducing than 
when it was increasing. Drain current will therefore 
be higher at given drain voltages, as those voltages 
are passed with increasing sweep voltage, than when 
passed with decreasing sweep voltage. In other 
words, the top part of the loops is traced first. 



Fig. 2-3. 

Drain breakdown 2N4416 junction FET 
voltage increases with increase in 
temperature. Drain current decreases 
with temperature. Compare Fig. 2-2. 



V G S = 0V 

10 20 30 40 50 

V QS ( VOLTS )- 

Fig. 2-4. Depletion mode operation showing 

breakdown. Insulated gate h'KT M-100. 


abrupt at 
lowI D 

See Fig. 2-4. The curves in this figure show a 
depletion mode, N-channel, insulated-gate field effect 
transistor operated in the depletion region between 
zero volts input bias and pinch-off. Breakdown is 
apparent for all amounts of gate voltage in that 
figure, but the breakdown region is most abrupt when 
the drain current is least. Fig. 2-5 shows the same 
transistor operated in the enhancement mode. The 
resistor used in series with the sweep voltage in the 
curve tracer limits the peak voltage appearing across 
the drain and source terminals at different values of 
drain current depending on the gate-drain bias. 

10 20 30 40 






Fig. 2-5. Enhancement mode operation same FET 
as in Fig. 2-4. 



Fig. 2-6. Zero bias drain current. Insulated 
gate FET M-100. 



The measurement of the forward transfer characteristics 
of field effect transistors is usually a matter of 
determining how much output current will flow at a 
given drain voltage with specific amounts of voltage 
at the input. Or the measurement may be to determine 
transconductance, how much change in output current 
there is with a specific change of input voltage. 

I DSS — Zero Bias Drain Current 

An important transfer characteristic is the value of 
output current which flows when the input voltage is 
zero. The symbol for this characteristic is IdSS — 
the drain current that flows when the source is 
shorted to the gate. Zero voltage exists between 
source and gate when the two are shorted together, 
of course. That drain current will depend on drain 
voltage too, so drain current should be measured at 
a specified drain voltage. Temperature may make a 
considerable difference, so temperature should also 
be specified. 

Fig. 2-6 shows the drain current which flows in a 
typical N-channel insulated gate field-effect 
transistor when the gate and source are shorted 
together and the drain voltage is swept between zero 
volts and 25 volts at room temperature. Fig. 2-5 
shows curves for the same transistor operated with 
gate voltage drive that starts at zero volts and 
changes in +0.5-volt steps to enhance (increase) 
drain current. However, this field-effect transistor 
is intended primarily to be operated with negative 
gate voltage drive, that is drive that depletes 
(reduces) drain current. Fig. 2-4 shows the forward 
transfer characteristics of the same transistor 
operated in its normal, depletion mode. 


Vt> or I'nc mvv^ — Pvnnh-nff Vnl.tnna 

A characteristic which is important for depletion- 
mode operation is pinch-off voltage — the input 
voltage required to turn off drain current 

Fig. 2-7 shows the characteristic curves for drain 
current versus gate voltage for a junction field 
effect transistor operated with the drain supply 



7 GS = -2.5V 


Fig. 2-7. Depletion mode operation of junction 

FET 2N4416. 


p inch-off 


I nanoamp 

voltage swept between zero and ten volts. Gate 
voltage was Increased negative in 0.5-volt steps 
into the pinch-off region. The figure shows very 
little current remaining with -2.5 volts applied 
between the gate and source terminals. Drain current 
might therefore be said to be pinched off at -2.5 
volts. However, if the vertical scale is expanded 
ten times (see Fig. 2-8), a current of a little less 
than 40 uA can be observed to be flowing with -2.5 
volts applied to the gate terminal. Pinch-off voltage 
for this field effect transistor is specified by 
the manufacturer to be less than 6 volts when the 
drain-to-source voltage is 15 volts, and the drain 
current is 1 nA or less. 


7 GS = " 2 - 5V 


Fig. 2-8. Pinch-off region 2N4416 junction FET. 



Some "eo n le will rnnsiHpr Hrain rnrrmit to be pinched 
off when it is less than 1% of the zero bias drain 
current that flows when the drain voltage is at some 
low value near the knee of the curves. For the set of 
curves in Fig. 2-7 the knee voltage would be about 
2 volts. Drain current at that drain voltage is about 
6 mA when the gate-to-source voltage is zero. Any 
drain current less than 60 yA (1% of 6 mA) might 
therefore be considered pinch-off current, similar 
to cutoff current for bipolar transistors. 

Fig. 2-9 shows the depletion mode characteristics 
of the same field effect transistor as in Fig. 2-7 
and Fig. 2-8, but with reduced drive and a different 

7 GS " ° V 

= -1.0V 

Fig. 2-9. 2N4416 junction FET. 7 GS source 
resistance In. Depletion mode 

control led 

control led 


vertical scale factor. The purpose of showing 
Fig. 2-9 is to make it easy to contrast depletion 
mode operation of this junction field effect transistor 
with enhancement mode operation. Fig. 2-10 shows the 
enhancement mode. A comparison of Fig. 2-10 with 
Fig. 2-9 shows a vast increase in drain current when 
the gate-to-source voltage is increased from +0.8 
volts to +1.0 volts. Below +0.6 volts the change 
in drain current with each equal increment (or 
decrement) of gate-to-source voltage is nearly equal. 
The reason for the radical change at about +0.6 volts 
is that the gate-source junction becomes forward 
biased, and the device starts to behave similar to 
a bipolar transistor: It starts to become current 
controlled instead of voltage controlled. The 

7 DS (VOLTS)- 



ii VW 

Fig. 2-10. 2N4416 junction FET. V GS source 
resistance 1Q. Enhancement mode 


gate supply 

iooo n 
vs I n 

-■ J -IT *-U« ~~*- n T. A 1f nnn n .. nn 1.r K.»,.,^rT»t!.f ITOV1T 

■LUiyCUCllH-C Ui Lilt g,ciO>- tUAUugi. o^f>i>-^_; ^wwwu.wv; '*-*-./ 

significant for enhancement mode operation of junction 
field effect transistors. To illustrate this point 
Fig. 2-11 and 2-12 show depletion versus enhancement 
operation of the same field effect transistor as in 
the former figures. There is a 2.5 to 1 greater 
voltage drive, but from a 1000-ohm supply resistance 
compared to a one-ohm supply resistance. Notice 
the great reduction of maximum drain current in 
Fig. 2-12 compared to Fig. 2-10. The cause, of 
course, is due to a larger than 2.5 to 1 reduction 
of gate-to-source voltage when forward current starts 
to flow. The reduced gate voltage is due to the IR 
drop across the 1000-ohm supply resistance. 

7 GS = ° V 

= -2.0V 


Fig. 2-11. 2N4416 depletion mode operation. 
F rs supply resistance = 1000Q. 


F„( SUPPLY) = +1.0V 


Vr.r. = OV 

2 4 6 8 10 



+ Q (W\ 










Fig. 2-12. 2N4416 junction FET. V GS supply 
resistance 1000Q. Enhancement 
mode operation. 


G — Transconductance (or transadmittanoe) 

The transconductance of a field effect transistor is 
an important forward transfer characteristic. It is 
an expression which tells us how much change in output 
current may be induced by a change in the input 

G = hi I LV , at a given drain-to-source voltage 
m u \jo 

Transconductance is typically higher in any region 
where drain current is high than in regions where 
G m increases drain current is low. Past the knee, drain voltage 
with J_ has only a slight effect on drain current for a given 
gate voltage. But the effect of drain voltage on 
transconductance is even less if the drain current is 
kept constant as the drain voltage is changed. In 
other words, for a given field effect transistor the 
transconductance is more a function of drain current 
than of drain voltage. 

Transconductance is proportional to the vertical 
distance between curves for a family of drain current 
versus drain voltage curves, such as shown in Fig. 2-7. 
In that figure, at a drain voltage of 5 volts, drain 
current drops from 7 mA (Point A) to 2.8 mA (Point B) 
for a change in gate voltage from zero volts to -1.0 
volts. That is a change of 4.2 mA in output current 
due to a one volt change in the input voltage. 

G m m '° 04 1 volt 1 " 68 - - 0042 mhos - 420 ° ymhos 

Fig. 2-13 shows an insulated-gate field effect 
transistor driven by a step generator which steps 
in one-volt increments from -2 volts to +2 volts. 
The transconductance for an input voltage near zero 
G m greater with 5 volts between drain and source is somewhat 
in greater for the enhancement direction than for the 

enhancement depletion direction. With +1 voit applied to the 
mode gate, drain current increases by 1.4 mA, so 

transconductance is 1400 micromhos. With -1 volt 
applied, current decreases by almost 1.1 mA so 
transconductance is a little less than 1100 
Tm'rromhos . 


V QS - + 2.0V 







Fig. 2-13. Insulated gate FET M-100 operated 
in both depletion and enhancement 
modes . 


1.16mA AT 10V 




— vw 









Fig. 3-1. Interbase resistance with emitter terminal 
open. 10 V/ .00116 I = 8600 n. 2N5061. 



Unijunction transistors are semiconductor devices 
having three terminals but only one junction. Two 
of the terminals are attached to opposite ends of a 
small bar of properly doped n-type semiconductor 
material. Somewhere along the bar a junction is 
made, and the third terminal is attached to the 
material that forms a junction on the bar. That 
terminal is called the emitter terminal and the other 
two are called base terminals - base one (B^) and 
base two (B£) • 

The conductivity of the bar is fairly linear, and 
^a can be measured with an ohmmeter by leaving the 

emitter terminal open. Or it may be measured on a 
transistor curve tracer. In Fig. 3-1 the interbase 
*£ o £ resistance was measured to be 8600 ohms with 10 

. volts applied. With zero voltage applied between 
^ the emitter terminal and B-^ the junction will be back 

| biased by the voltage drop V E in the bar between the 
*, ' junction and B-±. That voltage drop will depend on 

the amount of current flowing in the bar and the 
resistance of the bar material between the emitter 
terminal and B^. The voltage drop constitutes a 
back bias for the emitter junction. When enough 
voltage of the proper polarity is applied between 
the emitter terminal and base one the back bias will 
be overcome and forward current can flow across the 
emitter junction. When this happens carriers are 
injected into the bar material making it a better 
conductor, particularly in the region between the 
emitter terminal and Bj_. The voltage drop in that 
region, therefore, diminishes and as it does the 
emitter current suddenly increases because the 
original source of back bias diminishes. The effect 
negative is regenerative - the greater emitter current induces 
resistance greater emitter current. Unijunction transistors 
are principally used for this negative resistance 




3 TO B 1 (VOLTS) 

STEP. .47V 
TO B, 







Fig. 3-2. 2N4851 unijunction transistor. 

Their conductance characteristics can be explored 
using a transistor curve tracer. In Fig. 3-2 one of 
four values of current is applied to the B 2 terminal 
at all times. The currents are increased from zero 
to 60 uA in 20-uA steps 120 times a second. During 
the time each value of B 2 current flows, the emitter 
terminal is swept from zero volts to some value which 
causes 16 mA of emitter current to flow, then goes 
back to zero. The voltage that appears between the 
emitter terminal and the B-j terminal when the emitter 
sweep voltage is near zero, is caused by the voltage 


drop in the bar resulting from current between #2 an ^ 
B]_. That voltage drop is different for each 
incremental value of step current applied to the Bi 
terminal and can be easily identified by the bright 
dots spaced along the lower horizontal graticule 
line. For example, the third 20-yA step (60 pA) 
causes a voltage drop of .47 volts between the #2 
terminal and the B^ terminal. 

Each time the sweep voltage increases enough to turn 
the emitter on, the current flowing between the bases 
switches to a higher value and emitter current also 
switches to a higher value. See Fig. 3-3. 


S 2 TO B, (VOLTS) 

STEP; 1.5V 


B 2 TO B ] 





Fig. 3-3. 2N4851 unjunction transistor. 




Most of the conductance characteristics of four-layer 
semiconductor devices can be explored and measured 
on a transistor curve tracer. The characteristics of 
principal interest that may be measured are: 1) 
forward and reverse blocking (breakdown) voltages 
and currents; 2) the voltage drop at various forward 
currents for the on condition; 3) the gate-terminal 
turn-on voltage and current requirements for various 
values of applied anode-cathode voltage; 4) the value 
of forward current that holds the device in an on 
condition (holding current) . 

Thyristors are the same kind of semiconductor device 
as Silicon Controlled Rectifiers. The name thyristor 
is derived from thyratron, a gas tube controlled 
voltage- or rectifier. The name Silicon Controlled Rectifier 
current- is to distinguish the solid-state device from the 
operated gas-tube device. Thyristors are largely used to 
switch control the conduction duty factor of applied 

alternating voltage, but they do have many other 
applications. They can be turned on at any time the 
applied voltage is of the correct polarity but then 
cannot be turned off until the applied voltage 
approaches zero volts, or the current which is 
flowing diminishes to a value that is very low 
compared to the permissible peak value. 


The forward blocking voltage of a thyristor is the 
voltage that may be applied between cathode and anode 
before the device switches to have a low impedance — 
assuming little or no voltage or current is applied 
to the control terminal and that the polarity of the 
cathode-anode voltage is correct. 

Making a measurement of the forward blocking voltage 
of a thyristor using a curve tracer is done in very 
much the same way as measuring the reverse breakdown 
voltage of a transistor. First the gate terminal is 
usually either shorted to the cathode terminal or 


returned to the cathode through a resistor of 
specified value. In Fig. 4-1 forward blocking voltage 
was measured at 114 volts for 5 uA of forward current 
at room temperature (Point A) . The thyristor is rated 
to pass no more than 5 uA of peak forward blocking 
current at 60 volts, the rated peak forward blocking 
voltage, at a junction temperature of 125°C. A 
temperature-controlled oven would be needed to conduct 
the test at 125°C. Reverse blocking voltage would be 
measured in precisely the same way except the 
polarity of the sweep voltage would be reversed. 


40 80 120 160 200 

7 p (VOLTS)- 




Fig. 4-1. Forward blocking voltage and current 
2N5061 thyristor. 



Fig. 4-2 is similar to Fig. 4-1 except the applied 
voltage was increased until the thyristor switched 
to its on state with no gate voltage applied and 
a different vertical scale factor was used. A 
current-limiting resistor having a high resistance 
value was selected to limit the forward current. As 
the sweep voltage approaches its peak value and 
Point A is reached, avalanche breakdown occurs at the 
middle one of the three junctions, and the four-layer 
device appears to be simply two forward biased PN 
junctions in series. Current suddenly increases 

V f (VOLTS) 




Fig. 4-2. Switching conditions, 2N5061, with 
zero gate voltage drive. 


therefore, limited by the series resistance, and the 
forward voltage drop decreases to a very low 
value - Point B on the curve. As the sweep supply 
voltage increases further to its peak value, forward 
current increases from Point B to Point C. Current 
then diminishes as the sweep supply voltage drops 
toward zero. At Point D, not enough forward current 
remains to hold the thyristor in the on condition 
and current switches off to Point E. 

The value of current at Point D is the holding 
current for that set of conditions. The conditions 
existing for Fig. 4-2 are not a normal mode for 
operating a thyristor but represent a set of 
boundary conditions. Forward voltage is not usually 
applied if it exceeds the rated forward blocking 
voltage. And some current or voltage is usually 
applied to the gate terminal to switch the thyristor 
on. Fig. 4-3 is very similar to Fig. 4-2; the only 
difference is that a small steady value of turn-on 
voltage was applied to the gate terminal for Fig. 4-3. 
Two important differences should be noted: 
Switching takes place at a lower voltage and the 
value of holding current is reduced. 

Holding current is usually specified to be equal to 
or less than some maximum value under stated 
conditions of temperature, load resistance and anode 
supply voltage. To select the specified value of 
load resistance using a transistor curve tracer, both 
holding the value of the current-measuring resistor and the 
current selectable series resistor must be considered. 

Sometimes the correct value may be achieved only by 
using a third resistor applied at the test terminals. 
The gate voltage required to switch a thyristor to 
the on state at any given applied anode-cathode 
voltage can be determined on a Tektronix Type 576 
transistor curve tracer. By adjusting the peak supply 
voltage to the specified amount while the gate 
terminal voltage is zero the gate voltage can then be 
slowly increased until switching occurs and the gate 
voltage then read from the dial. Go-no go tests can 
be made by first dialing up the specified gate 
voltage and observing whether switching occurs or 
fails to occur. The source resistance for gate voltage 
drive may be specified. If so, the source resistance 
can be simulated by adding a resistor of appropriate 
value in series with the supply. 


C B D E 


40 80 120 160 200 

V ? (VOLTS) ► 




■ i WV- 

Fig. 4-3. Switching conditions, 2N5061, with 
small gate voltage drive. 


The gate current required to switch a thyristor to 
the on state may be tested or measured by means 
similar to those used for gate voltage turn-on 
measurements . 

Fig. 4-4 shows the high-current forward-conductance 
on characteristics of the same thyristor as used in 
the foregoing figures. The forward voltage drop at 
a current of one ampere is 1.3 volts. The specified 
maximum is 1.7 volts. 


V F 





v/m tacit 







Fig. 4-4. Forward conductance, 2N5061. 
Alternately on and off. 




Diodes present a much lower resistance to the flow of 
current in one direction than for flow in the opposite 
direction. Except for tunnel diodes and back diodes, 
simple PN junction diodes present less resistance to 
current flow when the P material is biased positive 
with respect to the N material, than when biased in 
the opposite direction. Tunnel diode and back, diode 
characteristics are discussed in a different section. 

If vs 7p The resistance that a diode presents to the flow of 
nonlinear a current in the forward direction is not linear 

over the entire operating range. That is, the forward 
current is not proportional to the forward voltage, 
although one increases when the other increases. 
A measurement of forward resistance, then, depends on 
the current and the voltage. 

If vs Vf For most purposes transistor curve tracers do a fine 

display job of showing the forward voltage and forward current 
relationships of a diode. The curve displayed is 
simply a graph of the forward current versus the 
forward voltage. The curve is typically produced 
by applying a variable (sweep) voltage of the correct 
polarity through a selectable resistor to one terminal 
while the other terminal is grounded. Then the voltage 
drop across the diode is applied through an amplifier 
to one set of deflection plates while the current 
through the diode is monitored through a series 
resistor and the voltage drop across the resistor 
applied through an amplifier to the other set of 
deflection plates. When the gain of the amplifiers 
is correctly set to match the deflection plate 
sensitivity of the cathode-ray tube, the horizontal 
and vertical scale will correspond to specific amounts 
of voltage and current. 

Simple DC instruments may also be used to measure 
current at specific voltages, or measure voltage-drop 
at specific currents. 


Vf for 
GaAs > 

Si > 

Similarly doped, and similar junction-area diodes made 
of germanium, silicon, or gallium arsenide will have 
considerably different forward characteristics. 
Small germanium diodes will have a forward voltage 
drop of about .4 volts, small silicon diodes about 
.8 volts and small gallium-arsenide diodes about 
1.2 volts at a given forward current of about 10 mA 
near the knee in the forward conduction curve. 
Fig. 5-1 is a triple exposure of the forward curves 
for three such diodes. 

Ge Si 


8 1.2 1.6 2.0 




— wv 


1 r 


vni 7AR 





— wv 


Fig. 5-1. Forward conductance of small 

germanium, silicon and gallium 
arsenide diodes. 


Jp vs V F 

Large junction-area diodes of a given semiconductor 
material will have knees at comparable forward voltage 
drops to small diodes but the curves may not be simple 
to compare. Large diodes offer less resistance to 
specific currents than small diodes made from the same 
material so comparable larger current scales should be 
used in larger diodes for such a comparison. 

Fig. 5-2 shows a double exposure of the forward 
voltage versus forward current curves of two silicon 
diodes that are rated for considerably different 



.4 .8 1.2 1.6 2.0 

V f (VOLTS) 




Fig. 5-2. Forward conductance characteristics 
of two silicon diodes at high 
currents . 


maximum forward currents. Fig. 5-3 shows the same 
two diodes using a different current scale. Note 
the greater similarity in the latter figure where 
forward current is lower. 

The measurement of forward DC resistance of the two 
diodes shown in the double exposure Fig. 5-2 at a 
forward voltage drop of .8 volts shows the small 
diode to be 16 ohms and the large diode to be 4 ohms. 










Fig. 5-3. Forward conductance characteristics 

of two silicon diodes at low currents. 


forward DC 


R = 


The forward DC resistance is simply the forward 
voltage (7 F ) divided by the forward current (Ip) 
at a specified voltage or current. The two 
measurements could have been made with DC meters. 

Differential resistance is measured between two 
points on a curve representing the forward current 
versus forward voltage. See Fig. 5-4. In that 
figure the points chosen were on the basis of a 
current difference of 50 milliamperes (from 50 mA 
to 100 mA) . The difference in forward voltage-drop 
for those two points was difficult to scale, so the 
horizontal deflection was magnified 10 times as shown 
in Fig. 5-5. The voltage difference can be seen to 
be 2.2 divisions, or 44 mV. 





Fig. 5-4. Forward resistance. 



.66 .70 




Fig. 5-5. Forward dynamic resistance of a small 
silicon diode between 50 mA and 100 mA. 

AFp = 44 mV; Alp = 50 mA. 
.044V / .050 I = .S8fi 

i imitations 

When the forward current, through a diode becomes 
excessive, the diode wiii heat up and fail. The 
temperature of the diode will rise as the forward 
current as increased, but the average temperature 
may not rise greatly if the average power does not 
rise greatly. The peak current that a diode can 
tolerate will depend on how long the peak current 
flows and on the duty factor of the applied peak 
current. When a very high peak current is applied 
it will cause a rapid internal temperature rise 


which may destroy the diode in a very short time 
the first time applied. Small diodes may be tested 
for relatively high forward currents using current 
pulses available on some transistor curve tracers. 


Voltage is reverse for all simple PN-j unction diodes, 
exoe-pt tunnel diodes and back diodes, when the 
polarity of the applied voltage is positive on the 
terminal that goes to the N material compared to the 
terminal that goes to the P material. 

The reverse current that flows when a diode is 
reverse biased depends on the reverse bias voltage. 
For most diodes the reverse current is relatively 
small up to the breakdown region. At and beyond this 
region small increases in voltage may cause large 
increases in current. Beyond the knee of a curve 
showing the transition from low reverse current the 
differential resistance may become fairly linear even 
though much lower in value. 

Reverse current is frequently called leakage current. 
However, the term leakage current is also used to 
distinguish a component of reverse current — that 
component which flows around the periphery of the 
principle junction area, including conduction through 
or over the surfaces of the diode case. 

Transistor curve tracers are excellent instruments 
for monitoring and measuring the reverse voltage 
characteristics of diodes under static conditions 
and low frequency dynamic conditions. Reverse 
static currents from less than 1 nanoampere (1 x 1Q-9 
measurements amperes) may be measured. Reverse voltage of 1000 

volts or more is also common. Reverse current versus 
reverse voltage is plotted on any convenient scale 
and measurements made at any point . 

Most diodes are rated to pass no more than a specified 
reverse current when a specified reverse voltage is 
applied. To verify such a specification using a 
transistor curve tracer the peak sweep voltage is 
manually increased to or beyond the voltage specified 
and current read from the scale. Usually a high value 
resistor is selected and switched in series with the 
non " diode and the sweep voltage. The resistor will limit 
destructive the current and protect the diode from excessive 
testing dissipation in the region of breakdown. Transistor 


1 vs Z> 2 

curve tracers proviue a luaiiuoi-xj-vaLioux^ f^ j - r 

voltage that occurs at the power line rate (usually 
50 or 60 hertz). The sweep voltages are selectable 
for either plus or minus polarity and are full-wave- 
rectified power line voltage. But the sweeps may 
also sometimes be alternately plus and minus — in 
which case they are unrectified versions of the power 
line voltage. Connecting two diodes in series and 
back-to-back while sweeping with alternate polarity 
voltage is a good way to compare or match the reverse 
characteristics of two diodes. See Fig. 5-6. In 
this figure the diodes are seen to have very similar 
reverse characteristics. In Fig. 5-7 one of the 
diodes was exchanged for another of the same type 
which differs considerably. 

Fig. 5-6, 


-V D (VOLTS)- 


Similar breakdown voltage and resistance 
of two diodes, type CD8204, placed back 
to back in series and swept with 
alternating voltage. 

Junction diodes that have been conducting in the 
forward direction do not instantly offer a high 
reverse resistance when the polarity of applied 
voltage is suddenly reversed. Current carriers which 
have become mobilized and diffused in the region of 
the junction due to forward current, will provide 



-100V +100V 
Fig. 5-7. Dissimilar breakdown characteristics 
of two diodes of same type as in 
Fig. 5-6. 

t i me : t 

measuri ng 


conduction In the reverse direction for a short 
while. Reverse current flows until the area can be 
depleted of carriers and become a depletion region. 
Whereas the carriers would eventually disappear due 
to recombination if the forward current simply 
stopped, their removal can be accelerated by applying 
reverse voltage. Removal by applying reverse voltage 
results in reverse current momentarily. How long the 
reverse current will flow depends on the reverse 
recovery time of the diode. But reverse recovery time 
will also depend on how much current was flowing 
originally. And it will depend on how much reverse 
current was induced to flow and how steadily that 
reverse current flowed while it lasted. The reverse 
recovery time of a diode is usually measured by 
applying a specific forward current through the 
diode, then periodically diverting that current with 
a current-step or voltage-step having a very short 
risetime and a specific amplitude sufficient to 
cause reverse current to flow momentarily. The 
length of time the reverse current flows will be the 
reverse recovery time of the diode. The current is 
usually monitored on an oscilloscope and the time it 
flows measured directly. To precisely define the 
time measurement, some point on the leading edge of 


the. applied turn-off pulse is defined as the 
beginning of the recovery time interval. This point 
is usually somewhere between 1% and 10% of the pulse 
amplitude. And some point on the trailing edge of 
the induced reverse-current pulse is then defined as 
the end of the recovery time interval. For 
convenience the beginning of the recovery time 
interval may also be defined as some point on the 
leading edge of the reverse-current pulse. Then the 
whole time interval can be measured using just one 
waveform. How the measurement should be made must 
be carefully described if reasonable correlation of 
measurements is expected. 

Included in such a description should be the amplitude 
of the pulses, and the source impedance and risetime 
of the pulse generator used to produce the pulses that 
divert forward current, and reverse bias the diodes. 
define pulse The oscilloscope response time and sensitivity must 
character also be known to be adequate. For measuring the 
reverse recovery time of fast diodes, sampling 
oscilloscopes are usually necessary. 

Just as important as the characteristics of the 
instruments to be used is the need to use very 
similar diode test fixtures — if good correlation 
of measurements is expected. Nearly any test 
fixture will show differences in recovery time 
between different diodes. But to accurately 
correlate measurements of the same kind made using 
different instruments and test fixtures is difficult. 
For measuring the reverse recovery time of the fastest 
diodes a test fixture should employ circuits 
carefully designed and constructed using transmission 
line techniques, to mask lumps of capacitance and 
inductance associated with discrete components. The 
diode itself must appear to be as nearly an integral 
part of a transmission line as possible. The circuit 
in Fig. 5-8 is such a diagram. That circuit is 
essentially the same one used for the photographs 
in Figs. 5-10 through 5-15. 

Fig. 5-9 shows the turn-off pulse applied to the 
diodes tested in the subsequent figures. The 
displayed risetime is approximately 0.6 nanoseconds 
and the amplitude is 600 mV when terminated in 50 
ohms. The picture was taken with a short, straight, 
bare wire used in place of a diode. A Tektronix Type 
111 Pulse Generator was applied to a Type 1S1 Sampling 
Plug- in unit through the test circuit. A Tektronix 
Type 549 Storage Oscilloscope was used for the display. 





COAX (oz 



Fig. 5-8. Circuit of reverse recovery- time 
test fixture. When terminated in 
50 ohms the output voltage is equal 
to 50 mV per ma of current. 

2 4 


Fig. 5-9. Turnoff pulse from Tektronix Type 111 
Pulse Generator. 


Figs. 5-10, 5-11, 5-12 and 5-13 are each photographs 
of two stored traces using four different diodes. 
The traces depicting low amplitude pulses in each 
figure correspond to a condition where no forward 
current was applied. The amplitude is a direct 
function of diode capacitance for those traces. The 
other four traces resulted from suddenly turning off 
2 mA of forward current and applying reverse voltage. 
The vertical scale is 2 mA per division. Forward 
current is shown in the top two divisions of the 
display and reverse current in the bottom four 
divisions. Let us assume reverse recovery time 
begins at the point corresponding to the vertical 
graticule line which is one major division from the 
left side of the scale. And let us assume the 
recovery time is said to end when the reverse current 
diminishes to 1 mA, one half of the forward current. 


I = 0mA 

2 4 

Fig. 5-10. Reverse recovery time, diode T13G. 


Fig. 5-10 does not permit us to make this measurement 
because the horizontal time scale of 1 ns per division 
is too short. We can say however, that the reverse 
current nearly reaches 8 mA and that it diminishes to 
2 mA in 8 ns. 

In Fig. 5-11 a different type diode was used. Reverse 
current reaches 6 mA, diminishes to 2 mA in about 2.8 
ns and recovers to 1 mA in about 5.8 ns. 


2 4 6 

Fig. 5-11. Reverse recovery time, diode GD238. 


In Fig. 5-12 a still faster diode was used. Reverse 
current goes to about 6 mA, recovers to 2 mA in a 
little over 1.2 ns and reaches 1 mA in about 1.4 ns. 
Notice that in spite of the shorter recovery time 
for the diode tested for Fig. 5-12, compared to the 
one tested for Fig. 5-11, its junction capacitance 
was greater. 

Fig. 5-13 shows very low diode capacitance and no 
reverse current except that due to diode capacitance. 
The diode used for this test was not a junction diode 
but a metal-contact, Schottky barrier diode. 


I = OmA 

2 4 

Fig. 5-12. Reverse recovery time, diode CD8204. 

2mA OmA 

2 4 6 


Fig. 5-13. Reverse recovery, Tektronix Schottky- 
barrier diode. 


Snap-Off uvoaes {anccp-uvoaea, Step-meovery uxoaesi 

the snap in 

The reverse recovery characteristics of snap-off 
diodes is unique. Whereas they may have a relatively 
high junction capacitance and a long reverse recovery 
time, the stored charge due to the presence of 
mobilized carriers disappears suddenly once the 
number of carriers is reduced sufficiently. That is 
the moment when the diodes "snap" or suddenly complete 
their recovery. 

Fig. 5-14 shows the reverse recovery characteristics 
of a snap-off diode with 2 mA, 1 mA and zero mA of 
forward current. The zero mA curve shows no storage, 
only reverse current due to the junction capacitance. 
The 1 mA curve shows a high reverse current that lasts 
about 2 ns. The 2 mA curve shows a considerably 
longer recovery time, but a comparable fast recovery 
once current starts to diminish. 

OmA 1 mA 2mA 

2 4 6 

Fig. 5-14. Reverse recovery, snap-off diode. 


Fig. 5-15 shows a similar set of curves for the same 
diode with current per division five times greater 
(10 mA/division compared to 2 mA/division) . The 
amplitude of the turn-off pulse was also increased 
five times (3 volts compared to 0.6 volts). Notice 
the similarity and the speed of turn-off once 
reverse current starts decreasing, regardless of the 
reverse recovery time. The three curves correspond 
to 10 mA, 5 mA and zero mA of forward current. 
Notice, incidentally, the reduced amplitude of the 
short pulse that corresponds to recovery from zero 
forward current, compared to Fig. 5-14. The reduced 
amplitude is partly attributable to the reduced 
capacitance of a more highly reverse-biased junction. 
It is also significant to note that even though the 
displayed final recovery time (snap-time) was 
comparable to the curves showing recovery from 2 mA 
and 1 mA, that the rise rate was approximately five 
times faster because of the increased amplitude. 

0mA 5mA 1 0mA 

2 4 6 

Fig. 5-15. Reverse recovery, snap-off diode. 






by different 
rneltiods not 

As one can see from the previous discussion of 
reverse recovery time, there are considerable 
differences in recovery time depending on how a diode 
is forward biased and how it is turned off. There is 
a need to describe and measure the characteristic of 
a diode that determines what its recovery time will 
be under various conditions. Looking analytically 
at a diode for the factors that limit its speed of 
recovery from conducting in the forward direction, 
one primarily sees mobilized carriers near the 
junction area that can turn around and provide a 
current path in the opposite direction. Until they 
all recross the junction, or recombine, reverse 
current can be high. All of these available carriers 
are similar to a charge stored in a relatively large, 
leaky capacitor. This capacitance, for the most 
part, is not real, although a small percentage of it 
may be. Once the mobile carriers all cross the 
junction, the real element of capacitance charges in 
the reverse direction to match the applied reverse 
voltage. That element of capacitance diminishes 
somewhat as reverse voltage builds up. The real 
part of junction capacitance typically has very little 
to do with recovery from forward current. It does 
contribute considerably to momentary reverse current, 
but essentially to the same extent regardless of the 
amount of forward current. In terms of recovery from 
forward current, then, only stored charge is 

Measurement of stored charge seems to be a more direct 
way of determining the comparative merits of a diode 
as a fast switch. Of course junction capacitance has 
a bearing on the switching speed of a diode also, but 
that can be measured separately. 

Instruments are made which indicate stored charge 
directly. The principle of the operation of 
insti-uments having circuits which follow the JF.mC 
Suggested Standard No. 1 (June, 19CC) is basically 
one of averaging the area under repetitive reverse 
current pulses. The problem of correlation between 
instruments of different types using different test 
circuits, or circuit layouts, can still exist, 
however. Different test circuits may see a different 
area under each current pulse of fast diodes if the 
circuit construction differs substantially. The same 
average of different areas will yield different 


Fig. 5-16. 


Knowledge of the performance characteristics of a 
diode as a high frequency rectifier is often required. 
Junction capacitance and carrier storage are principal 
limiting factors for such an application. A simple 
basic circuit may be used, such as in Fig. 5-16, 
typical of a high-frequency detector circuit. The 
ratio of the output DC voltage to the peak value of 
the applied sinusoidal voltage will be an indication 
of the diode's rectification efficiency. 

In a way this method of testing the high frequency 
characteristics of a diode is similar to measuring 
stored charge. The amplitude of the applied voltage, 
the impedance of the sine-wave generator, the size of 
the load capacitor and the time constant of the 
output load will have a bearing on the measured 
efficiency. Great care must be given to stray 
capacitance and lead inductance in this kind of 
circuit for the faster diodes. 


The junction capacitance of a diode is usually 
measured with a capacitance bridge. Measurements 
usually intentionally include the stray capacitance 
bridge for of the package and leads. However, the added 
junction C capacitance of long leads may be neutralized at the 
bridge by using the right kind of test fixture. 
When capacitance is measured without back bias being 
applied to the junction, the test signal amplitude 
should be no more than about 100 mV peak to prevent 
the diode from conducting. 


Back bias may be applied to measure the junction 

f" qn qr" 1 t" anr*o a t" rlifforonf i7a1itoo <-\f roiioi'fo 17^1 t'Qno 

Junction capacitance usually decreases as reverse 
bias is increased and the depletion region widens. 
Varaator diodes are made and used for their voltage- 
variable capacitance characteristics. 


The forward recovery time of most diodes is much less 

t. vs t than the reverse recovery Lime. There is lit Lie Lo 

f r rr 

recover from except stored charge in the junction 

capacitance, if the diode is reverse biased. However, 
even without a previously applied reverse bias, some 
diodes will take a considerable time to become a good 
conductor in the forward direction. A given forward 
current will momentarily cause a high forward voltage- 
drop. After the minority carriers become fully 
concentrated in the area of the junction, current 
flows with less resistance. Some factors which 
contribute to the fast recovery from forward current, 
impede the mobilization of carriers for turn-on. So 
some diodes with fast reverse recovery may have 
relatively poor forward recovery characteristics. 

Forward recovery time may be measured in much the 
same way as reverse-recovery time: Time is measured 
from some point on the rise of a turn-on pulse to 
some point on the curve near the end of the 
momentarily excessive voltage drop across the diode. 
Ideally a constant current would be applied in the 
forward direction and the voltage drop across the 
diode then monitored. The fastest measurements of 
diode turn-on time, like those of reverse recovery 
time, require the diode to be in a transmission-line 
environment. This generally precludes measurement of 
the voltage directly across the diode. Diode drive, 
fortunately, does not have to be from a strictly 
constant-current source for the change in forward 
voltage-drop of the diode to be monitored. Turn-on 
time may be measured and compared using the same, or 
similar, circuits as employed for measurement of 
reverse-recovery time. 



V — Zener Voltage, 7. . 

Zener diodes are designed to be used in the reverse 
voltage breakdown mode. It is normal for them to 
conduct in the high impedance (reverse) direction. 
When they are conducting in the forward direction 
they behave very much the same as ordinary signal 
diodes or rectifier diodes. The principle use for 
voltage zener diodes is to provide a nearly constant 
reference voltage drop. In the breakdown voltage region the 
dynamic, or differential, resistance is relatively 
low, so that a considerable change in reverse current 
may occur while the zener voltage remains nearly 

measure Zener voltage can easily be measured using a current 
with meter source and DC meters, or it may be measured using a 
transistor curve tracer. When using a transistor 
curve tracer, an appropriate value of resistance is 
placed in series with the sweep voltage supply, then 
the peak sweep voltage increased slowly until the 
zener voltage is reached and the desired amount of 
peak reverse current is observed to flow. Zener 
diodes are made in various sizes with widely 
different wattage ratings. The maximum reverse 
current will depend on how much temperature rise 
may be tolerated. Ideally there is no reverse 
current until the zener voltage is reached. 


Fig. 6-1 shows a Tektronix Type 576 Transistor Curve 
Tracer plotting the reverse voltage versus reverse 
current of a zener diode. The zener voltage can be 
seen to be approximately 6.2 volts. A more precise 
measurement of zener voltage may be made by using 
the ten times magnifier to expand the horizontal 
deflection, then using the calibrated offset 
(positioning) control to bring the region back near 
center-screen. Fig. 6-2 shows the zener voltage to 
be 6.22 volts when the reverse current is 25 mA. 








Fig. 6-1. Zener voltage, V(BR) » 1N753A. 


5.5 5.7 5.9 6.1 6.3 6.5 



+ jrm 







Fig. 6-2. Zener voltage, V rgm , and 

differential resistance, Z ZT , 


The effect of temperature on zener voltage may be 
easily observed on a curve tracer. Usually it is 
only necessary to increase the peak reverse current 
zener sufficiently. The temperature will increase as 
temperature current is increased, and this usually causes zener 
limits voltage to increase, for zener voltages above 5 or 
6 volts. Some zener diode packages are made which 
have a very low temperature coefficient. 



70 72 74 76 78 80 

V a (VOLTS)- 



( <7 

V *1 


— vw 



< i.i rr dc 

Fig. 6 3. Effect of temperature on zener voltage. 
Double exposure of same 75-V zener 
diode. Extra peak current uf curve B 
causes increase in temperature and 
increase in breakdown voltage. 


Fig. 6-3 is a double exposure photograph showing a 
zener voltage of 74.6 at a reverse current of 2.5 mA 
(Point A) when the peak current is 2.9 mA. When the 
peak current is increased to 4.7 mA the zener 
voltage increases to 75.3 volts at the same reverse 
current. See Curve B. 

Z — Zener Impedance, DC 

The DC impedance (resistance) of a zener diode 
operated in the breakdown-voltage region will depend 
on the voltage and the current. By measuring the 
precise voltage at a given current, then dividing 
that voltage by the current, the zener impedance, 
Z v , may be calculated. 

Z„ T — Zener Impedance, AC 

The dynamic, or differential impedance (resistance) , 

of a zener diode is revealed by the slope of the 

current-versus-voltage curve in the zener region. 

This curve is typically very steep, depicting a low 

differential resistance, when the current is plotted 

on the vertical axis. To measure differential 

resistance in this region, two points on the curve 

are chosen which bracket the area of prime interest, 

and the voltage and current at these two points then 

noted. Differential resistance will be the quotient 

of the difference in the two voltage points, A7(BR) , 

and the difference in the two current points, AJ B . 


Z ZT = A7 (BR) /AJ R 

In Fig. 6-2 voltage increases approximately one minor 
division (20 mV) when current increases from 10 mA 
to 25 mA (15 mA) . 

Z = 20/15, or approximately 1.3 fl. 

Whenever the curve is a fairly straight line, 
choosing two widely separated points that bracket the 
area of prime interest will improve the accuracy of 
the measurement. If the curve is not nearly 
straight, a small straight-edge carefully held 
tangent to the point of prime interest will allow 
points to be chosen on the scale which are more widely 
separated, and thereby improve the accuracy of the 
measurement . 




Tunnel diodes are used primarily for their negative 
negative resistance characteristics. They make very fast 
resistance low-power switches and oscillators. Their very fast 
switching capabilities make it difficult for 
curve tracers to plot the negative resistance 
portions of their current-versus-voltage curves 
without very special attention being given to the 
diode holder design. Lead inductance and stray 
capacitance are usually high enough to cause the 
diode to oscillate while operated in its negative 
resistance region, regardless of what DC load line 
is used. But for most purposes it is sufficient to 
know the limits of the negative resistance region, 
and curve tracers show these limits quite well. 

Back diodes are essentially the same kind of devices 
as tunnel diodes, but their use does not generally 
depend on their having a negative resistance 
characteristic. They are like very-low-voltage 
rectifiers, and may in fact be called tunnel 
rectifiers. Both tunnel diodes and back diodes 
present less resistance to current flow when the 
PN-junction is biased with the P-material negative 
with respect to the N-material than when biased in 
forward? the opposite direction. This is contrary to other 
back? diodes, and poses the question of which direction 
should be considered the forward direction. The 
term tunnel direction will be used in this discussion 
when the applied bias makes the P-material positive 
with respect to the N-material. That polarity is 
correct to cause tunnel diodes to tunnel, and back 
diodes to tunnel, or at least tend to tunnel. It is 
the direction of higher resistance to current flow. 


rigs, /-i anu i—i. snow uue uuuiiex uj.recuj.on 
characteristics of two tunnel diodes having a 20 to 
1 difference in peak point currents. 

I p and V p 

J v and F v 




Fig. 7-1. Tunnel -direction conductance curves 
for 100 -mA tunnel diode. 









Fig. 7-2. Tunnel-direction conductance curves 
for 5-mA tunnel diode. 


Fig. 7-3 is a double exposure showing both the 
tunnel diode curve of Fig. / — *■ anu tue ior«aiu 
conduction curve of an ordinary small silicon diode. 






,i -wv- 

Fig. 7-3. Tunnel direction conductance of 5-mA 
tunnel diode (curve A) compared to 
forward conductance of small silicon 
diode (curve B) . 


Figs. 7-4 and 7-5 show the tunnel direction curves 
of a back diode. 




Fig. 7-4. Tunnel-direction conductance of 
BD-4 back diode. 







Fig. 7- 5. Tunnel -direction conductance of same 
back diode as in Fig. 7-4; different 
vertical scale. 


Fig. 7-6 is a bi-polarity curve showing both the 
tunnel direction curve of a tunnel diode, and the 
low impedance direction curve of that tunnel diode. 






Fig. 7-6. Tunnel-direction and opposite 

direction conductance of 5-mA tunnel 
diode. Note lower impedance for 
opposite direction conduction. 





Fig. 7-7. Bi-polarity conductance of back 
diode compared to tunnel diode; 
double exposure. 

Fig. 7-7 Is a double exposure showing bi-polarity 
curves of a back diode and a tunnel diode. 

Fig, 7-1 shows the tunnel direction conductance 
characteristics of a 100-mA tunnel diode. The 
following currents and voltages can be read from 
this figure: 

Peak point current 0.1 amps 

Valley point current 0.01 amps 

Peak point voltage 0.14 volts 

Projected peak point voltage 0.61 volts 

Valley point voltage U.44 volts 



The peak-point current of a tunnel diode is the 
maximum value of current that may flow at a low value 
of applied voltage. Currents above the peak-point 
current value are beyond the negative resistance 
region, so correspond to only one possible value of 
voltage. The projected peak-point voltage is the 
lowest value of voltage that may correspond to a 
current in excess of the peak point current. 

The peak point voltage of tunnel diodes is the 
highest voltage that may exist across a tunnel diode 
without exceeding the peak point current. The peak 
point is easily recognized in either Fig. 7-1 or Fig. 
7-2. It is the top point on the left hand section 
of the curves. 


The valley point is the lowest point on the curve 
beyond the peak point. This point is sometimes a 
little more difficult to determine precisely than 
the peak point because the slope may be nearly 
horizontal for a considerable distance. Once 
determined, the valley point current and the valley 
point voltage can be read directly from the scale. 
In Fig. 7-2 the valley point voltage is somewhere 
between 400 and 450 millivolts. Peak points will 
always be visible on a curve tracer, however; they 
will not be somewhere in the negative resistance 


The dynamic or differential resistance of a tunnel 
diode or back diode depends on the particular diode, 
and how it is biased or operated. Differential 
differential resistance is a function of the slope of the 
R _ AK conductance curve. It can be measured by choosing 
two points on a curve, scaling the difference in 
current and in voltage between those points, and 
dividing the difference in voltage (&V) by the 
difference in current (AJ). Where a slope is 
changing, a straight edge held tangent to the point 
of interest will identify the slope. Any two points 
on the straight edge may then be chosen to determine 
the differential resistance. 




Negative resistance could be measured from the slope 
of the curve in the negative resistance region if we 
were to show a curve in that region. However, the 
average slope in that region is fairly close to the 
slope of an imaginary line connecting the peak point 
and the valley point. The resistance of a curve 
with that slope can be determined from the difference 
between the valley point voltage and the peak point 
voltage and the difference between the peak point 
current and the valley point current: 

In Figure 7-1: 

Resistance = T 

y = .44 volts 

V = .14 volts 

J = .10 amperes 

J = .01 amperes 

So the average negative resistance is 
equal to: 

(.44) - (.14) volts _ 30 = 3 3 a 
(.10) - (.01) amperes .09 

The difference between valley-point voltage and peak- 
point voltage is between about 300 and 400 millivolts 
for all tunnel diodes. Therefore the negative 
resistance typically varies inversely with the peak 
point current of different tunnel diodes. 



The definitions of the listed terms are the same as 
appear in Publication 147-0 by the International 
Electrotechnical Commission, 1966, except for the 
deletion of some notes that pertain to a term or 



General terms 

Terminal (of a semiconductor device) 

A specified externally available point of connexion. 

Electrode (of a semiconductor device J 

That part providing the electrical contact between the specified region of a semiconductor device 
and the lead to its terminal. 

Forward direction (of a PN junction) 

The direction of continuous (direct) current flow in which a PN junction has the lowest resistance. 

Note. — This definition may not apply to tunnel devices. 
Reverse direction (of a PN junction) 

The direction of continuous (direct) current flow in which a PN junction has the higher resistance. 

Note. — This definition may not apply to (unnel devices. 

Physical terms 


A material with resistivity usually in the range between metals and insulators, in which the 
electrical charge carrier concentration increases with increasing temperature over some temperature 

Extrinsic semiconductor 

A semiconductor with charge carrier concentration dependent upon impurities or other imperfec- 

N-type semiconductor 
Extrinsic semiconductor in which the conduction electron density exceeds the mobile hole density. 

P-type semiconductor 
Extrinsic semiconductor in which the mobile hole density exceeds the conduction electron density. 

/-type (intrinsic) semiconductor 

Nearly pure and ideal semiconductor in which the electron and hole densities are nearly equal 
under conditions of thermal equilibrium. 


A region of transition between semiconducting regions of different electrical properties. 
P N junction 

A junction between P and N type semiconductor material. 
Alloyed junction 

A junction formed by alloying one or more materials to a semiconductor crystal. 
Diffused junction 

A junction formed by the diffusion of an impurity within a semiconductor crystal. 


Grown junction 
A junction produced during the growth of a semiconductor crystal from a melt. 

Charge carrier (abbreviation: carrier) 
In a semiconductor, a mobile (free) conduction electron or mobile hole. 

Majority carrier (in a semiconductor region) 

The type of carrier constituting more than half of the total charge carrier concentration. 
Minority carrier (in a semiconductor region) 

The type of carrier constituting less than half of the total charge carrier concentration. 

Depletion layer 

A region in which the mobile charge carrier density is insufficient to neutralize the net fixed charge 
density of donors and acceptors. 

Breakdown (of a reverse-biased PN junction) 

A phenomenon, the initiation of which is obseived as a transition from a state of high dynamic 
resistance to a state of substantially lower dynamic resistance for increasing magnitude of reverse 

Avalanche breakdown (of a semiconductor PN junction) 

A breakdown that is caused by the cumulative multiplication of free charge carriers in a semi- 
conductor under the action of a strong electric field which causes some free carriers to gain enough 
energy to liberate new hole-electron pairs by ionization. 

Avalanche voltage 
The applied voltage at which avalanche breakdown occurs. 

Thermal breakdown (of a semiconductor PN junction) 

A breakdown that is caused by the generation of free charge carriers owing to the cumulative 
interaction between increasing power dissipation and increasing junction temperature. 

Note. — This effect is also known as thermal runaway in some countries. 

Zener breakdown (of a semiconductor PN junction) 

A breakdown caused by the transition of electrons from the valence band to the conduction band 
due to tunnel action under the influence of a strong electric field. 

Zener voltage 
The applied voltage at which Zener breakdown occurs. 

Tunnel effect (repetition of definition 07-16-015 of 1EC Publication 50 (Q?) **ut omitting !..e note, 
The piercing of a potential hill by a carrier, which would be impossible according to classical 
mechanics, but the probability of which is not zero according to wave mechanics, it the widlli c.l 
the hill is small enough. The wave associated with the carrier is almost totally reflected on the first 
slope, but a small fraction crosses the hill 

: Tunnel action (in a PN junction) 

A process whereby conduction occurs through the potential barrier due to the tunnel effect and 
in which electrons pass in either direction between the conduction band in the N-region and the 
valence band in the P-region. 

Nate. — Tunnel action, unlike the diffusion or charge carriers, involves electrons only and for all practical purposes 
the transit time is negligible. 


Photo-electric effect 

Interaction between radiation and matter resulting in the absorption of photons and the consequent 
generation of mobile charge carriers. 

Photovoltaic effect 
A photo-electric effect wherein an electromotive force is generated. 

Types of device 

Semiconductor device 

A device whose essential characteristics are due to the flow of charge carriers within a semi- 

Semiconductor diode 
A two-terminal semiconductor device having an asymmetrical voltage-current characteristic. 

Note. — Unless otherwise qualified, this term usually means a device with the voltage-current characteristic typical 
of a single PN junction. 

Voltage reference diode 

A diode which develops across its terminals a reference voltage of specified accuracy, when 
biased to operate within a specified current range. 

Voltage regulator diode 

A diode which develops across its terminals an essentially constant voltage throughout a specified 
current range. 

Semiconductor rectifier diode 

A semiconductor diode designed for rectification and including its associated mounting and 
cooling attachments if integral with it. 

Tunnel diode 

A diode having a PN junction in which tunnel action occurs giving rise to negative differential 
conductance in a certain range of the forward direction of the current-voltage characteristic. 


A semiconductor device capable of providing power amplification and having three or more 

Note. — Other names may be used to describe certain special types of semiconductor device covered by this definition. 

Photoconduciive cell 

A device in which the photoconductive effect is utilized. 
Photovoltaic cell 

A device in which the photovoltaic effect is utilized. 

A diode in which the photoelectric effect is utilized. 

A transistor in which the photoelectric effect is utilized. 


Terms related to ratings and characteristics 

Reverse voltage 

The voltage across a junction or a diode when biased in the direction corresponding to the higher 

Note. — This definition may not apply to tunnel diodes. 

Floating voltage 

Voltage between an open-circuited terminal and the reference point when a specified voltage 
is applied to any of the other terminals. 

Breakdown voltage 

Reverse voltage at which the reverse current through a junction becomes greater than a specified 

Cut-off frequency 

The frequency at which the modulus of a measured parameter has decreased to l/|/2 of its low 
frequency value. 

Note. - For a transistor, the cut-oH trequency usually applies to the shurl-wivuil mall-signal fur*ard current transfer 
ratio for either the common-base or common-emitter configuration. 


Case temperature 
The temperature measured at a specified point on the case of a semiconductor device. 

Thermal derating factor 

The factor by which the power dissipation rating must be reduced with increase of ambient or 
case temperature. 


Continuous (direct) reverse voltage 

The value uf the constant voltage applied to a diode in the reverse direction 

Peak reverse vo!iag> 

The highest insU 
repetitive and non-repetitive transients. 

T-. i ■ , . ■ , . . ...,i..- ->f .U„ r~ ^cc KoltiiTf AT.irritw nrrms a diode including all 

The highest insUiiUiituun vuldv o! ...... ...,....„. .-..-. e - ~ ^ 

Reverse current 

The total conductive current flowing through the diode when specified reverse voltage is applied. 

Forward voltage 

—, . ,. , ._• _ .. i_ ...v:_t. „ it" f-~ m <Kp fln...' nfi> in the forward direction. 

ihe voltage acius* ific Iciiiniitus "jiivii ioui.j a^m <••- ■*- '•■ ~> --^ ---.-■ 

The current flowing through the diode in the direction of lower resistance. 


Differential resistance 

The differential resistance measured between the terminals of the diode under specified condi- 
tions of measurement. 

Forward d.c. resistance 

The quotient of d.c. forward voltage across the diode and the corresponding d.c. forward 

Reverse d.c. resistance 

The quotient of the d.c. reverse voltage across the diode and the corresponding d.c. reverse 

Small-signal capacitance 
Differential capacitance at the diode terminals, measured under given bias conditions. 

Reverse recovery time 

The time required for current or voltage to recover to a specified value after instantaneous 
switching from a specified forward current condition to a specified reverse bias condition. 


Forward direction 

The direction of current flow within the diode for which the characteristic includes negative 
differential conductance. 

Reverse direction 

The direction of current flow within the diode for which the characteristic includes only 
positive differential conductance. 

Peak point 

The point on the characteristic corresponding to the lowest voltage in the forward direction 
for which the differential conductance is zero. 

Peak point voltage 
The voltage value at the peak point. 

Peak point current 
The current value at the peak point. 

Valley point 

The point on the characteristic corresponding to the lowest voltage greater than the peak point 
voltage for which the differential conductance is zero. 

Valley point voltage 
The voltage value at the valley point. 

Valley point current 
The current value at the valley point. 


Peak to valley point current ratio 

The ratio of the peak point current to the valley point current. 
Projected peak point 

The point on the characteristic where the current is equal to the peak point current, but where 
the voltage is greater than the valley point voltage. 

Projected peak point voltage 
The voltage value at the projected peak point. 

Peak point current 

Projecte d peak point voltage 
Valley point voltagB 

Peak point voltage 

Static voltage-current characteristic of a tunnel diode. 
Negative dijferenlial conductance region 

That part on the characteristic of a tunnel diode between the peak and valley points. 
Case capacitance 

The residual capacitance between the device terminals when the PN i 

connected . 

I junction is not internally 

Series inductance 
The total effective internal series inductance under specified conditions. 


General terms 

Base terminal 
The specified extrrnaltv available point of connection to the h > 


Collector terminal 
The specified externally available point of connection to the collector region. 

Emitter terminal 
The specified externally available point of connection to the emitter region. 

Collector region 
A region between the collector junction and the collector electrode of a transistor. 

Emitter region 
A region between the emitter junction and the emitter electrode of a transistor. 

Base region 
A region between the emitter and collector junctions of a transistor. 

Collector junction 

A junction between the base and collector regions normally biased in the reverse direction and 
through which the charge carriers flow from a region in which they are minority carriers to one 
in which they are majority carriers. 

Emitter junction 

A junction between the base and emitter regions normally biased in the forward direction, and 
through which the charge carriers flow from a region in which they are majority carriers to one 
in which they are minority carriers. 

Types of transistor 

Junction transistor 

Transistor having a base region and two or more junctions. 

Note. — The operation of a junction transistor depends upon the injection of minority carriers into the base 

Bi-directional transistor 

A transistor which has substantially the same electrical characteristics when the terminals 
normally designated as emitter and collector are interchanged. 

Note. — Bi-directional transistors are sometimes called symmetrical transistors. This term, however, is deprecated 
as it might give the incorrect impression of an ideally symmetrical transistor. 

Tetrode transistor 

A four-electrode transistor, usually a conventional junction transistor having two separate 
base electrodes and two base terminals. 

Unipolar transistor 
A transistor which utilizes charge carriers of only one polarity. 

Terms related to ratings and characteristics 

Punch-through voltage 

The value of the collector-base voltage above which the open-circuit emitter-base voltage 
increases almost linearly with increasing collector-base voltage. 

Note* — <"Jleach-through voltage" is a term also used in the U.S.A. 


saturation vottage 

The residual voltage between collector and emitter terminals under specified conditions of 
base current and collector current, the collector current being limited by the external circuit. 

Cut-off current (reverse current) 

Reverse current of the base-collector junction (or base-emitter junction) when the emitter 
or the collector is open-circuited, the reverse voltage being specified. 

Emitter series resistance 

The resistance between the emitter terminal and the internal inaccessible emitter point in an 

equivalent circuit. 

Saturation resistance 

The resistance between collector and emitter terminals under specified conditions of base 
current and collector current, when the collector current is limited by the external circuit. 

Note. The saturation resistance may be determined either as the ratio of total voltage to total current or as the 

ratio of differential voltage to differential current ; the method of determination must be specified. 

Extrinsic base resistance 

The resistance between the base terminal and the internal inaccessible base point in an equi- 
valent circuit. 

Collector depletion layer capacitance 

The part ofthe capacitance across a collector-base junction that is associated with its depletion 

Note. — The collector depletion layer capacitance is a function ofthe total potential difference across the depletion 

Delay time (of a switching transistor) 

The time interval between the application at the input terminals of a pulse which is switching 
the transistor from a non-conducting to a conducting state, and the appearance at the output 
terminals ofthe pulse induced by the charge carriers. 

Note. — The time is usually measured between points corresponding to 10% of the amplitude of the applied 
pulse and of the output pulse, respectively (see Figure 1 , page 5 1 ). 

Rise time (of a switching transistor) 

The time interval between the instants at which the magnitude of the pulse at the output 
terminals reaches specified lower and upper limits respectively when the transistor is being 
switched from its non-conducting to its conducting state. 

Note. — The lower and upper limits are usually 1 0% and 90°; respectively, of the amplitude of the output pulse. 
(see Figuic 1). 

Carrier storage time (of a switching transistor) 

The time interval bcUixn the U&iiiuJiig of thv fall of the puis* applied to Liu; input lei mmal» 
and the beginning ofthe fall ofthe pulse generated by charge carriers at the output terminals. 

Note. -— I he time is generally measured between the 90% values of the two pulse amplitudes (sec Figure 1). 

Fall time (of a switching transistor) 

The time interval between the ln?:t:\ntf at which the magnitude of the nnjrc at the out n ut 
terminals reaches specified upper and lower limits respectively when the transistor is being switched 
from its conducting to its non-conducting state. 

Note. The upper and lower limits are usually 90:„ and 10?; respectively, of the amplitude uf the output pulse 


Small-signal short-circuit forward current transfer ratio 

The ratio between the alternating output current and the small sinusoidal input current 
producing it under small-signal conditions, the output being short-circuited to a.c. 

10% -. 


= delay time 
= rise time 

H = fai 
h = cai 

rier storage time 

Applied pulse 
(idealised wave shape) 

Output pulse 
{idealised wave shape) 

Output pulse 
{practical wave shape) 

Switching transistor pulse characteristic. 

Static value of the forward current transfer ratio 

The ratio between the continuous (direct) output and the continuous (direct) input current, 
the output voltage being held constant. 

Small-signal - open-circuit reverse voltage transfer ratio 

The ratio of the alternating voltage appearing at the input terminals, when they are a.c. open- 
circuited, to thealternating voltage applied to the output terminals, under small-signal conditions. 

Circuit configuration 

Common base 

Circuit configuration in which the base terminal is common to the input circuit and to the 
output circuit and in which the input terminal is the emitter terminal and the output terminal is the 
collector terminal. 


Inverse common base 

Circuit configuration in which the base terminal is common to the input circuit and to the 
output circuit and in which the input terminal is the collector terminal and the output terminal 
is the emitter terminal. 

Common collector 

Circuit configuration in which the collector terminal is common to the input circuit and to the 
output circuit and in which the input terminal is the base terminal and the output terminal is the 
emitter terminal. 

Inverse common collector 

Circuit configuration in which the collector terminal is common to the input circuit and to the 
output circuit and in which the input terminal is the emitter terminal and the output terminal 
is the base terminal. 

Common emitter 

Circuit configuration in which the emitter terminal is common to the input circuit and to the 
output circuit and in which the input terminal is the base terminal and the output terminal is the 
collector terminal 

Inverse common emitter 

Circuit configuration in which the emitter terminal is common to the input circuit and to the 
output circuit and in which the input terminal is the collector terminal and the output terminal 
is the base terminal. 



Except as noted, the symbols and definitions of 
symbols are the same as in Electronic Industries 
Association Standard RS-245A. 


B7 CB0 Obsolete - see F (BR)CBQ 

BV CEQ Obsolete - see 7 (BR)CEQ 

57 CER Obsolete - see ^ (BR)CER 

BV QES Obsolete - see 7 (BR)CES 

BV rvv Obsolete - see V ,„„.„„„ 


BV-m>n Obsolete - see V , 

EBO ° c "(BR)EBO 

C ±^ Open-circuit input capacitance, common base 

C ±l >s Short-circuit input capacitance, common base 

C ± eo Open-circuit input capacitance, common emitter 

C j_ es Short-circuit input capacitance, common emitter 

C \ )0 Open-circuit output capacitance, common base 

C \y S Short-circuit output capacitance, common base 

C 0&o Open-circuit output capacitance, common emitter 

C oes Short-circuit output capacitance, common emitter 

■^hfb Small-signal short-circuit forward current transfer 
ratio cutoff frequency (common base) 

•fhfe Small-signal short-circuit forward current transfer 
ratio cutoff frequency (common emitter) 

•f T Frequency at which small-signal forward current 

transfer ratio (common emitter) extrapolates to 










Static forward current Lrausiei' ratio vCOuuuOn base) 

h Small-signal short-circuit forward current transfer 

ratio (common base) 

h Static forward current transfer ratio (common 


h Small-signal short-circuit forward current transfer 

ratio (common collector) 

h Static forward current transfer ratio (common 

FE . .. .. 


h Small-signal short-circuit forward current transfer 

ratio (common emitter) 

h Static input resistance (common base) 


h.. Small-signal short-circuit input impedance (common 


h Static input resistance (common collector) 


h Small-signal short-circuit input impedance (common 


h Static input resistance (common emitter) 

h Small-signal short-circuit input impedance (common 


h , Small-signal open-circuit output admittance 

(common base) 

h Small-signal open-circuit output admittance 

(common collector) 

h Small-Ri gnal open-circuit output admittance 

(common emitter) 

h , Small-signal open-circuit reverse voltage transfer 

ratio (common base) 

h Small-signal open-circuit reverse voltage transfer 

'" c ratio (common collector) 

h Small-signal open-circuit reverse voltage transfer 

e ratio (common emitter) 


If, Base current, DC 

? r Collector current, DC 

J CB0 Collector cutoff current, DC, emitter open 

-T CE0 Collector cutoff current, DC, base open 

J CER Collector cutoff current, DC, with specified 
resistance between base and emitter 

J CEV Collector cutoff current, DC, with specified 
voltage between base and emitter 

I CES Collector cutoff current, DC, with base short 

circuited to emitter 

-^DSS Drain current, DC, with gate shorted to emitter* 

-Z~ E Emitter current, DC 

J EB0 Emitter cutoff current (DC), collector open 

Jggg Gate leakage current* 

r CE ( sat ) Collector-to-emitter saturation resistance 

*Not a part of Standard RS-245-A 


Vf \nr>r\ Breakdown VOiLage, UUJ-letLul-LU-Uiiac, cuatui uptu 


v, . Breakdown voltage, collector-to-emltter , base open 
(BR) CEO 6 

K, „ Breakdown voltage, collector-to-emitter, with 




specified resistance between base and emitter 

y . Breakdown voltage, collector-to-emitter, with base 

short-circuited to emitter 

V. , Breakdown voltage, collector-to-emitter, with 

specified circuit between base and emitter 

|f, m ,.„ Breakdown voltage, drain-to-gate, source open* 

F . Breakdown voltage, drain-to-source, gate shorted to 
(BR)DSS 4. 


V, n „\T,™ Breakdown voltage, emitter-to-base, collector open 


1/ . Breakdown voltage, gate-to-source, drain shorted to 


y Base-to-collector voltage, DC 


7 Base-to-emitter voltage, DC 

V Collector-to-base voltage, DC 

V Collector-to-emitter voltage, DC 


v Collector-to-emitter voltage, DC with base open* 


V nccTic-N Collector-to-emitter (breakdown) sustaining 
CEO (SUS) i ^ j, 


V. , , Collector-to-emitter voltage, DC with specified 
resistor between base and emitter 

V Collector-to-emitter (breakdown) sustaining 
CER(SUS) voltage * 

V Coiiector-to-emitter voltage, DC wiLli base olioiL- 

circuited to emitter 

*Not a part of Standard RS-245-A 



CES(SUS) Collector-to-emitter (breakdown) sustaining voltage* 


7 EB 


7 GS 

Collector-to-emitter voltage, DC with specified 
circuit between base and emitter 

^CEX(SUS) Collector-to-emitter (breakdown) sustaining voltage* 
^ CE (sat) Collector-to-emitter saturation voltage, DC 
^ DS Drain-to-source voltage, DC 

Emitter-to-base voltage, DC 

Emitter-to-collector voltage, DC 

Gate-to-source voltage, DC 
^ RT Reach-through voltage 

*Not a part of Standard RS-245-A 



Admittance, output, 54, 62 
Alpha, 22, 23 
Avalanche breakdown, 39 
Avalanche breakdown, thy r is tor, 

Avalanche transistor, 41 
Back biased, 47 
Back diodes, 125 
Beta, 19-22 
Beta, AC, 15 
Beta, DC, 1, 3 

Bidirectional transistors, 15 
Bipolar transistor, 1, 56 
Bipolarity conductance curve, 

Breakdown, FET, 71-75 
Breakdown, reach through, 42 
Breakdown region, 29-32 
Capacitance, diode junction, 

Capacitance, virtual, 53 
Circuit, switching time, 52 
Collector breakdown, 37-38 
Collector sustaining voltage, 

Collector voltage saturation, 

Carrier diffusion time, 54 
Carrier recombination time, 54 
Carrier storage time, 44, 48 
Conductance curves, tunnel, 

Conductance diode forward, 

Conductance thyristor, 98 
Conduction time, 13 
Current gain, 1 
Current gain, high frequency, 

Current gain, unity, 55 
Curves, interpreting, 18-19 
Cutoff current, 28, 33, 35 
Cutoff frequency, 54-55 
Cutoff pulse, 52 

Definition, symbols , 147 
Definition, terms, 135 
Delay time, 44-47 
Depletion/enhancement mode, 87 
Depletion Mode, FET, 76-84 
Differential resistance, 103 
Differential resistance, tunnel 

diode, 133 
Differential resistance, zener, 

119, 121-123 
Diffusion time, 47 
Diode comparison, 106 
Diode, current vs voltage, 99 
Diode, forward conductance, 102 
Diode, forward resistance, 102-104 
Diode junction capacitance, 

Diode measurements , pulse 

characteristics, 108 
Diode, temperature effects, 104 
Diodes, Ge, GaAs, Si, 100 
Diodes, signal, 99 
Diodes, varactor, 118 
Drain current vs temperature, 75 
Drain current, zero bias, 78-79 
Duty factor, 6 
Dynamic resistance, 39 
Dynamic resistance, forward, 

Enhancement mode, FET, 77, 83, 

Emitter follower conductance, 

Fall time, 44-45, 52 
Field effect transistor, 71 
Forward bias , 32 
Forward blocking voltage, 93-94 
Forward conductance, diode, 

Forward current transfer, 1, 

13, 21 
Forward recovery time measure- 
ment, 118 
Forward resistance, diode, 102-104 


TTnrw^rH transfer. FET . 79 
Gain bandwidth product, 55 
Go, no-go, 4 

Go, no-go test, thyristor, 96 
Heat flow vs temperature, 7 
Heat symptoms, 8 
Holding current, 96 
Input capacitance, 48, 53 
Input impedance, 56-61 
Input resistance, 56-61 
Impedance, zener, 123 
Interbase, resistance, 88-89 
Junction capacitance, diode, 

Junction capacitance, tran- 
sistor, 48 
Junction temperature, tran- 
sistor, 5-13 
Leakage current, diode, 105 
Leakage current, FET, 71-73 
Linear operation, 20 
Loop , 8 

Measurement errors , 36 
Negative resistance, 36, 40, 
89, 125, 134 

Output capacitance, 54 

Output admittance, 62-67 

Parasitics, 41 

Peak power, 13 

Pinch-off specification, 82 

Pinch-off voltage, 80 

Power dissipated, 27 

Punch through, 42-43 

Reach through, 42 

Recovery time, forward, 118 

Rectification efficiency, 117 

Repetition rate, 10 

Recovery time, reverse, 106-113 

Resistance, saturation, 62 

Response delay, 50 

Reverse bias, 32 

Reverse blocking voltage, 9 5-94 

Reverse current, 28-29 

Reverse current vs temperature, 

Reverse current, high, 114-116 

Reverse recovery, snap-off, 

Reverse voltage, diode, 105, 

Reverse terminals, 14 

Reverse voltage transfer, 

Rise rate, 47-48 
Risetime, 44 
Saturation region, 24-27 
SCR, 93 

Silicon controlled rectifier, 93 
Single family, 5 
Single shot, 5 
Snap-off diodes, 114 
Small signal, 18 
Static measurements, diode, 105 
Step-recovery diode, 114 
Storage time, 50, 52 
Stored charge, 116 
Sustaining voltage, 39 
Switching time, transistors, 

Symbols, defined, 147 
Symmetrical transistors, 15 
Temperature effects, diode, 104 
Temperature effects, FET, 74-75 
Temperature effects, transistor, 

5-13, 30-31 
Temperature effect, zener, 122 
Terms, defined, 135 
Test fixture, reverse recovery 

time, 109 
Thyristor forward conductance, 98 
Thyristor switching, 95-97 
Thyristors, 93 
Transadmittance, FET, 86 
Transconductance, FET, 86 
Transistor, high-speed, 51 
Transistor loading effect, 56 
Tunnel diode conductance, 

Tunnel diode, peak point, 133 
Tunnel diodes, 125 
Tunnel rectifier, 125 
Turn-off characteristics, 44, 

Turn-off pulse, 109 
Turn-on characteristics, 44, 46 
Unijunction conductance, 90-91 
Unijunction transistors, 89 
Varactor diodes, 118 
Voltage breakdown, 28 
Zener diodes, 119 
Zener voltage, 119-121 


This book is but one of a series. 
The series consists of two groups, 
airauits and measurements. These 
texts present a conceptual approach 
to circuits and measurements which 
apply to Tektronix products. 
Several "concept books" are in 
preparation; those now available 
are : 

Power Supply Circuits 

Oscilloscope Cathode-Ray Tubes 

Storage Cathode-Ray Tubes and Circuits 

Television Waveform Processing Circuits 

Information Display Concepts